by Rohit Balkishan
This amplifier was born out of a need to use two sets headphones with my computer’s sound-card. The design presented here is a 50mW power amplifier meant for phones with impedances of 32 Ohms and greater. I chose this class A topology, because it offers very good distortion figures without a lot of complexity. A simple common emitter amplifier, for example, is not very linear and the overall gain is very much device dependent. In the case of my amp, it uses a voltage feedback (VFB) topology, and the gain is dependent only on the ratio of 2 resistors. Plus, the amp has very good power supply and common mode rejection on account of the differential input pair and the current source used to bias it.
With a simple common emitter amplifier, the gain is given by the formula (assuming hfe >> 1) Av = hfe * Rc / [(hfe * Re + hie)]. hfe is the transistor current gain and hie is its base-to-emitter resistance. Rc is the collector (or load) resistor which biases the transistor and Re is the emitter resistor which provides bias stabilization and local -ve feedback. hie is device dependent and highly non-linear. It varies with the collector (or emitter) current and causes the gain to vary with the collector current, resulting in distortion. If Re is large enough to make hie negligible, then Rc will also need to be large and the amplifier won’t be able to source/sink current into low-impedance loads.
My headphone amplifier is a conventional VFB type employing commonly available parts. Let’s consider a typical VFB setup using BJT transistors. We have a differential pair input stage, a voltage gain stage, an output (current gain) stage and a -ve feedback network. For the explanation, I will not include the current gain stage since it has no role to play as far as the voltage gain is concerned. The voltage gain stage is a CE stage with a constant current source (CCS) for the collector resistor and theoretically has an infinite voltage gain as per the above formula (the output impedance of a true CCS is infinite). Also note that the presence of a CCS minimises the variation in gain due to hie, which becomes extremely small and can be neglected, and also makes the use of an emitter resistor unnecessary.
Although the non-linearity aspect of the CE stage is minimized, the infinite (or very large) gain needs to be brought down to a more useful level. This is where the differential input stage and -ve feedback network come in. The differential stage operates by way of comparing the signals between it’s inverting and non-inverting inputs and tries to make the difference equal to zero. The signal to be amplified is applied to the non-inverting input (this is determined by where the CE stage gets it’s input from) and the -ve feedback network applies a part of the output of the CE stage to the inverting input.
Since the differential pair inputs are actually the bases of the transistors forming the differential pair, both are high impedance inputs and for practical purposes we can neglect the currents in them, if offset voltages are not an issue. Due to the infinite open loop gain, and the fact the the voltage difference between the inverting and non-inverting inputs is always zero, the closed loop gain is only dependent on the -ve feedback network, which in this case is simply a voltage divider connected across the output of the amp, and the feedback voltage taken from the point where the resistors are connected to each other.
Another thing to be noted is that due to the VFB topology, any change in temperature or supply voltage appears as a common-mode signal at the amp’s inputs and is suppressed (of course the temperature range must reasonably within the operating range of the individual devices). So this type of amplifier has an extremely stable gain which can be controlled simply with two resistors. Note that the VFB amp (without the feedback network – this is external) is in fact a discrete form of an op-amp, though not as good as an IC op-amp.
In the circuit in figure 1, Q1 and Q2 form the differential pair input stage with Q8 and Q9 as active loads. Current source Q3 biases the input stage at about 520uA. Q4 is the gain stage, biased at about 520uA by the current source Q5.
The output stage is comprised of the compound emitter follower Q6/Q10 and the compound current source Q7/Q11 which biases the emitter follower at about 124mA, which is about 10% higher than the peak output current (110mA, measured at the transistor side of C5). Q7/Q11 form the CCS along with D1 and R11/R12. This amp does not use any push-pull arrangement for the output stage.
CCS operation: The voltage accross R11/R12 is the zener diode voltage less the Vbe drop of Q7 (about 2V). This means that if the diode voltage is constant, the current thru’ R11/R12 will be constant => current in the emitter terminal of compound pair Q7/Q11 is constant => since, Ic = Ie (neglecting base current) the current in the load (Q6/Q10) connected to the CCS will be constant. R5 provides the diode current. Note that in the prototype, I have used LEDs in place of zeners. This is fine, except that a slight reduction in the o/p stage bias current will occur and the LEDs must be connected in a direction opposite to that of the zener (pointing downwards instead of upwards, in the schematic).
The closed-loop gain (excluding the input attenuator formed by R1, R2 and R3) is 1 + R7 / R6, as it is used as in the non-inverting mode. Here, R6 and R7 set the gain at 12dB. The gain is measured as the ratio of voltages (RMS) at the collector of Q10 and the base of Q1. R1 and R3 form an attenuator to decrease the input sensitivity of the amplifier to about 850mV for an output power of 50mW. To increase the sensitivity, R1 can be reduced. Changing the gain of the amplifier is possible by changing the feedback resistors R6 and R7. In the case of my amp, the feedback resistors have already fixed the gain to a value higher than needed and the input signal is attenuated to compensate for this. So to change the overall gain either the attenuator resistors can be changed or the -ve feedback resistors can be changed. If the attenuator is changed then the input caps must also be changed to maintain the lower and higher -3db frequency points.
D1 (2.7V, 0.25W) is the single voltage reference used by all three current sources (this can be replaced with a standard red LED, with a small reduction in bias currents). R5 sets the zener current to about 7mA. Q10 dissipates about 530mW and Q11 dissipates about 400mW and these need to be mounted on small heat-sinks.
The amp’s output impedance, Zo (as seen looking into R14 or R15, with both phones connected [assumed to be resistive, not inductive]), is 5.6 Ohms at frequencies > 100Hz, rising to 6 Ohms at 20Hz. This is as indicated by my simulator.
The ±5V power supply is a typical dual power topology (figure 2). Filtering is provided by the 2200uF/25V electrolytic capacitors. The voltage available at the caps is applied to the regulators as shown and the outputs of these are bypassed by 100nF ceramic caps. Note that the regulators don’t have any reverse voltage protection, since the PSU is meant to be assembled as part of the amplifier without any provision for connecting/disconnecting with the amp. Assembly is again not very critical except that the regulators must be fed from the caps and not the bridge and insulation must be proper to avoid shock hazards.
Here are the specifications (as simulated in SIMetrix):
Power: 2 x 50mW, 32 Ohm headphones.
Distortion: < 0.5%, 20 Hz to 20 KHz.
Input sensitivity: 850mV for rated power.
Frequency response (-1 dB): 5.5 Hz – 159 KHz.
The amp uses commonly available parts and can be constructed at a very low cost. It cost me about Rs. 200 (INR) (that’s about 4$ US) including everything (PSU, perf board, jacks, box, volume control, components, etc.). Note that this particular design is meant to be used as a distribution amp for 2 sets of 32 Ohm headphones of 50mW power each. Construction is not at all critical. I simply wired the circuit as it appears in the schematic. It is extremely stable and will not oscillate unless there is something seriously wrong with the wiring. I have mounted the components on a perforated board and connected them using ordinary stranded wires.
All resistors must be 1/4W, and 1% metal film types for minimum noise. The volume control, which is a 47K dual audio pot, is connected in a conventional manner as a variable voltage divider (wiper going to the amp’s input).
C2, C3 and C5 must ideally be non-polar types (rated at least 12V) although standard polarised electrolytics can be used. C2 and C3 can be obtained as non-polar units by simply connecting like poles of two 100uF capacitors (+ to + or – to -). The overall voltage and capacitance of the combined pair will be Ctot = 1 / (1 / C1 + 1 / C2), and the voltage rating will be the sum of the voltage ratings of the individual caps. For this amp though, polarised caps can be used without any problems since the voltage across the caps is < 50mV.
All electrolytics must preferably be bypassed with 100nF capacitors (not shown in the schematic). The 100nF caps are ceramic types, but film caps may be used (I really doubt if it’ll have any audible improvement though). C5 is used to prevent any DC from reaching the phones in the event of o/p device failure. Although a failure is very improbable, I have included it as a fail-safe, and can be omitted if a small amount of DC offset is acceptable. With C5 in place, the lower -3dB frequency becomes about 3 Hz with both A and B phones connected (1.8 Hz with either A or B). If C5 if not used, then C2 may be chosen to get a lower -3dB frequency of about 3 to 5 Hz. The amplifier may be used with headphones having an impedance of 32 Ohms or higher.
The transistors need not be very critically matched. For the power devices, the range of hfes allowable is 50 to the maximum available for BDxxx devices and the matching has to be done to ensure that hfes are within 20% to 30% of each other. As for the BCxxx devices the same above rules apply. The minimum hfe being 100. It may be noted that the output stage is a compound pair and as such, the combined hfe (of Q6/Q10 and Q7/Q11) is very high, making the absolute values and matching less critical.
Q10 and Q11 (optionally) need a heatsink of about 40mm x 40mm or thereabouts of 1 or 1.5 mm thick aluminum plate. These can be avoided if the box is big enough to provide good ventilation. The transistors dissipate about 600mW each, without a heatsink. So using heat-sinks will assure complete reliability. The prototype does not use heatsinks for the transistors.
It is very much possible to substitute transistors. The following is the min. required rating for the transistors Q1 to Q9 (BCxxx ratings in brackets):
Power: >= 0.1W (0.5W)
Ic: >= 20mA (100mA)
hfe (DC): >= 80 (150 typ.)
The min. required rating for Q10 and Q11 is (BDxxx ratings in brackets):
Power: >= 3W (8W)
Ic: >= 500mA (1.5A)
hfe (DC): >= 50 (100 typ.)
Looking at these ratings, I guess the choice of transistors for substitution is quite large, but the transistors that I have used provide a very wide safety margin and I would recommend that any substitutions be with equivalents rather than based on the min. required ratings. Of course, transistors with better ratings than BCxxx & BDxxx can be substituted without any problems.
The power supply must be ±5V regulated with a minimum current rating of 300mA per channel (600mA for stereo operation). It uses a 7.5-0-7.5VAC, 600mA transformer to step down the mains to a usable level. The rectifier bridge is comprised of 1N4007 diodes. Any IC bridge can be used instead of the diodes as long as it can handle up to 1A of current. The voltage regulators are mounted on heatsinks.
I haven’t used a fuse with the PSU of the amp that I have built. In spite of this, I would recommend using a 300mA fuse for 240V (600mA for 120V) mains in series with the transformer’s primary (use the nearest standard values available).
The picture above shows the internals of the amp. At the left is the power transformer. The PCB mounts, the PSU rectifiers, capacitors and regulators (on heatsinks) are next. The output caps can be seen towards the top edge of the PCB (one in the middle and the other towards the right edge). Just below the output caps are the output transistors (not on heatsinks) and their current source transistors and LEDs. The headphone out jacks and volume control pot are towards the bottom of the image. The power switch and input jack are towards the right-most end of the box. The input wires (from the input jack to the volume control) – these need to be of shielded type so that any noise pick-up is avoided.
I have used a plastic enclosure, since it’s very easy to work with. A metal enclosure can well be used, provided it’s properly earthed with care taken to avoid a ground loop. There is no hum or noise in the amp. By now it will be quite clear that I’m not very good at mechanical workmanship. The extra holes that can be seen in the front (to the left of the volume knob) and side (to the right of the power switch) views are a result of this.
As for the sound, I found it to be of the same quality as the source used to drive it, such as my computer’s sound card headphone out, which can also drive headphones directly. As may be noted, the amp has an attenuator in its input that makes it work with an overall gain of just above unity. As for comparison with other systems, the sound is better than the headphone output of my Kenwood portable music system, when both the headphone amp and the system are being fed by my Sony CD walkman.
The amp does not “improve” the sound of an existing source to which it is connected. It only gives a sound that is the same as that of the source. It’s not possible for a good amp to improve the sound of a bad source, but it is very much possible for a bad amp to spoil the sound of a good source. When the source is the PC or CDP, the amp’s sound is very good, and the amp doesn’t change it except by way of allowing a greater power output and to connect 2 sets of phones. As for the Kenwood system, the headphone output (built-in) is not at all comparable to either the PC’s output or my CD player’s output – it is noisier with quite some amount of audible hiss. So, in all, if I use my Kenwood system with an external source (CDP or PC), the sound is not as good compared to the output of my headphone amp being driven by the same CDP (and PC) for the same music.
Modifications for Driving High Z Headphones like the Sennheiser HD600
The Sennheiser HD600 is a high impedance headphone (300 ohms). As per the headphone power requirements table given in Dennis Bohn’s article, the max. power handling of the HD600 is 80mW. The modifications required are as follows (I have only simulated the circuit for 300 Ohm operation):
a) R7 = 39K. To increase the gain so that the voltage swing will be about ±6.5V.
b) R11 and R12 = 56 Ohms, 1/4W each (or a single 27 Ohm, 1/2W resistor. A 1/4W resistor can be used for the single 27-Ohm but will cause the resistor to heat up slightly). Bias current is reduced. DO NOT skip this step; otherwise the output devices will be running at a bias current that’s much more than needed, causing unnecessary heating.
c) The supply voltage needs to be ±10V DC regulated, at 300mA. This is to avoid clipping. Change the 7805 and 7905 regulators to 7810 and 7910 ones for the HD600 version. Also, the power transformer needs to have a secondary rated at 12-0-12V/300mA to cater to the 10V regulators. The rest of the PSU need not be changed.
d) R5 = 680 Ohms (this is optional). To keep the zener diode current nearly the same as that of the 32 Ohms version.
e) For the sake of reliability, Q10 and Q11 must be mounted on small heatsinks (each will dissipate about 650mW). Although the dissipation has not changed a lot from the 32 Ohm version, still I am suggesting the use of heat-sinks as I have only simulated the circuit for 300-Ohm operation.
The HD600 version must not be used with a low impedance phone as it is. The HD600 version can provide a larger voltage swing but less current. This is fine for the impedance and power requirements of HD600. The 32 Ohm headphone will (assuming a desired output of 50mW) will need greater current, and if connected will cause clipping (and possible overheating).
To make the amp work with both headphone types, the bias current can be increased to that required by 32 Ohm phones and the gain resistor R7 can be switched to different values depending on headphone impedance. To increase the bias current R11 and R12 need to have the older value of 33 Ohms each and R7 needs to be: 39K for 300 Ohm operation 10K for 32 Ohm operation Note that this will result in a dissipation of about 1.3W in the output devices, so good heat-sinking IS A MUST. The design basically remains that for 300 Ohm operation but the output stage is then biased at a higher current. So, the power supply needs to be ±10V at 300mA.
One way to achieve the gain switching would be to have R7 as a parallel combination of 2 resistors R7A and R7B such that – R7A is a 39K resistor and R7B is connected across R7A via a switch (figure 3). R7B needs to be about 15K. Close the switch for 32 Ohms and open it for 300 Ohms operation.
c. 2003 Rohit Balkishan.