by Chu Moy
Back in 1999, I received an email with an attached schematic of a tube amplifier. The sender told me that it was supposed to be the schematic for a “clone” of the famous EarMax miniature headphone amplifier. Since he had no more information about the design and had not built it, I filed it away to be referred to at a later date. Months later, I saw the schematic again in a book called Valve Amplifiers (2nd ed.) by Morgan Jones. In the book, Jones described it as a reverse-engineered version of the EarMax. That is, the schematic was not of the true EarMax, but was derived from the published specifications of the EarMax (e.g., 3 tubes, the power supply voltage, the input and output impedances). Jones had created the circuit as an academic exercise, but had not actually built it. I once again put the schematic away, hoping later to find a DIYer who could give construction details.
While doing research on the internet, I came upon a reference to the Jones design in an archive for the Sound Practices mailing list. Lance Dow, who knew Morgan Jones personally, had posted the schematic in that newsgroup way back in 1996. Dow had not built the circuit either. Given all the interest in the audiophile community about the EarMax, I thought that surely someone, somewhere, must have tried to build it. I scoured the Sound Practices archives, downloading year after year of digests, and finally found a posting by Johannes Chiu, who described enthusiastically his DIY work on this design. I contacted Chiu for more information, but apparently he no longer remembers many specifics about it. This previous version of this article was a summary of the information collected from the Sound Practices archives and from the HeadWize forums about the original Morgan Jones circuit.
Since the publication of that article over a year ago, there have been several inquiries regarding the possible topology of the EarMax Pro. The Pro has basically the same output tube complement as the EarMax, but can provide higher current output into low impedance headphones. The specifications suggested that the topology of the EarMax Pro was similar to the EarMax (both having White cathode follower output stages), but the mystery remained as to how two amplifiers with similar output stages (no output transformers) could have substantially different output characteristics. Then in 2002, Alex Cavalli submitted revised Morgan Jones circuits with new parts values, based on the White cathode follower optimization techniques developed by John Broskie. The optimized Morgan Jones amplifiers (with and without feedback) can output more than 3 times the current of the original into a 32-ohm load. The amp shown at the top of this article is an optimized Morgan Jones amplifier (without feedback) built by Bryan Ngiam.
The Amplifier Designs
1. The Original Morgan Jones Amplifier
Figure 1 is the schematic for the original Morgan Jones amplifier. It has a grounded cathode input stage with an idling of about 3mA. The output stage is a push-pull White follower, which provides low output impedance without the need for global feedback. It idles at about 10mA and can swing ±20mA in push-pull. The output impedance is about 10 ohms (the calculated value was 6 ohms). The overall gain of the amp is about 22 (the calculated value was 28). Jones used an ECC88 input tube. The original EarMax has an ECC81 input tube, but he felt that the low anode current required for this stage would lead to noise and gain problems with the ECC81. The EarMax output tubes are ECC86. Jones used the similar ECC88, later used in the EarMax Pro’s output as well. He rated the amplifier to drive headphones from 200 to 2000 ohms.
In all of following voltage and Fourier analysis graphs, the red curve is the input; the green curve is the output. The top graph in figure 2 shows the output voltage waveform into a 300-ohm load (the input is a 0.15V, 1 KHz sine wave). The bottom graph is a Fourier analysis of the output waveform to determine the harmonic distortion, which turns out to be about 2%. The output current (not shown) is 9mA, so the amplifier is driving the load with 36mW (the maximum power into 300 ohms is about 120mW).
Figure 3 shows the same set of graphs for a 32-ohm load, with a 0.016V, 1 KHz sine wave. Like the example in figure 2, the output is being driven to a total harmonic distortion of 2%, but here, there is a distinct imbalance between the top and bottom halves of the waveforms, because the lower impedance load draws more current and is unbalancing the push-pull configuration. The amplifier is supplying about 2.6mW into 32 ohms (the maximum power into 32 ohms is about 13mW). Thus, like the EarMax, the original Morgan Jones amplifier is not truly suited to power low impedance headphones such as the Grados, despite the low output impedance.
2. Analysis of the Performance of the Original Morgan Jones Amplifier
Theoretically, this type of output stage should be able to drive low impedance loads well, because it has a very low output impedance. In his article The White Cathode Follower, TubeCad editor John Broskie investigated the poor performance of the White cathode follower when driving low impedance loads. He discovered that the voltage drop across the anode load resistor (R4) of the top tube (V2a) varies with the current flowing through the tube. If the voltage across R4 is high enough, it will overdrive the bottom tube (V2b).
Alex Cavalli viewed the problem another way: the imbalance was caused by the gain of the V2a being greater than 1:
Assume that the output of the amplifier is shorted (an AC short at the junction of the upper and lower triodes in the output stage) and ignore the fact that the tubes are in series. Under this condition, both the upper and lower triodes are operating as simple grounded cathode amplifiers, where the output of the upper section is fed directly into the lower section.
Now the gain of the first stage is about 25. The gain of the upper triode in grounded cathode mode is about 15. If a 0.01V sine wave is applied to the input, the first stage will produce about 25 x 0.01 =.25V at its plate. Thus the upper triode sees 0.25V on its grid. The upper stage in turn produces 15 x 0.25 = 3.75V, which is coupled to the grid of the lower triode.
There are two things to note here:
- the grid drive to the push-pull sections is unequal, 0.25V (upper) vs. 3.75V(lower) and
- the bottom triode, which has a bias of about 1.75V, is being driven hard into cutoff and positive grid.
In this design the upper and lower output sections are not working together equally and so they are not producing the maximum possible current swing. Furthermore, the enormous gain feeding the lower section makes the amp extremely sensitive and sends it out of class A mode quickly.
The maximum voltage that can appear at the grid of the bottom tube is determined by the DC biasing voltage across the cathode resistor. In figure 1, for example, the bias or idling voltage across R5 is 1.7V, so the maximum peak-to-peak voltage into the grid of V2b is 1.7V. A higher grid voltage will either turn the tube off or drive the grid positive and will push the triode out of Class A operation.
Each graph in figure 5 shows the differential voltages between the grid and cathode (Vgk) of the output tubes V2a (blue) and V2b (magenta) in the original Morgan Jones amplifier driving a 300-ohm load. In the top graph, Vgkfor V2b is 1.66Vp-p, which is just less than the 1.7V bias voltage across R5. The Vgk for the top and bottom tubes appear symmetrical, but have unequal amplitudes. At this Vgk, the amplifier is outputting 2V into a 300-ohm load, which is approximately 13mW. This is the maximum output power of the original MJ amplifier into 300 ohms before push-pull stage leaves class A mode. The Fourier analysis indicates that the harmonic distortion at 13mW is only 0.3% (output voltage and Fourier graphs omitted).
The bottom graph shows the Vgk waveforms when the White follower stage is severely unbalanced. The Vgk for V2b exceeds 1.7Vp-p and the shapes of both waveforms are grossly distorted. Here, the original MJ amplifier is driving a 300-ohm load with an output voltage of 3.3V (about 36mW) and the harmonic distortion has risen to 2% (output voltage and Fourier graphs omitted).
After employing a similar White follower balance analysis for a 32-ohm load, the maximum output power of the original MJ amplifier into that load is actually less what was determined from figure 3: a mere 1.6mW at 1.3% distortion (0.228V output). Even for high efficiency, 32-ohm headphones like the Grados, 1.6mW is not enough to achieve clean volume levels – especially not if the music has wide dynamic range.
Broskie concluded that in order for the White follower to perform optimally (and maintain the balance in the push-pull pair), the anode load resistor R4 should be chosen so that the bottom tube receives an identical grid-to-cathode voltage signal as the top tube. In other words, Vgk for V2a should equal Vgk for V2b (the bias voltage across the cathode resistor still determines the limit of Vgk). His solution was to calculate a lower value for the anode load resistor (which he called Ra) based on the equation:
Ra = rp/mu = 1/Gm
where Gm is the transconductance of the tube.
Alex Cavalli provided this explanation:
The way to balance the grid drives where the output is shorted (a load of zero ohms) is to ensure that the upper output section has a gain of 1. This will cause the lower triode to see exactly the same grid signal as the upper triode. According to Broskie the effective Gm of the upper stage is:
Gm = (mu + 1) / (rp + Ra)
where Ra is the anode load resistor and rp is the plate resistance of the triode. The gain of upper stage is given by:
Gain = ((mu + 1) x Ra) / (rp + Ra)
To have a gain of 1:
((mu + 1) x Ra) = (rp + Ra)
Then solving for Ra:
Ra = rp/(mu) = 1/Gm
This result is the same as Broskie’s, except that he proves this result for all load impedances.
The transconductance for a 6DJ8 is 11mA/V, so Ra ~ 90 ohms.
3. The Optimized Morgan Jones Amplifier
Alex Cavalli’s revised Morgan Jones circuit is shown in figure 4. It is identical the original, except that the power supply’s current rating has been doubled and three resistor values in the amplifier have been changed. R2, R4, and R5 determine the balance for the White push-pull output stage. R2 determines the quiescent plate voltage on V1 which sets the grid bias on the V2a in combination with R4 and R5. These seemingly minor changes in the resistor values have a huge impact on the performance of the amp as discussed below.
He used PSpice simulations to determine the best values for R2, R4 and R5. Although Broskie determined that the optimal anode load resistor value R4 was 1/Gm (or 90 ohms for a 6DJ8), the simulations indicated that the amplifier had better output characteristics with a higher value – 150 ohms. The output stage still idles at around 10mA. These modifications resulted in better performance into both 300-ohm and 32-ohm loads.
The output voltage of the Cavalli-optimized amplifier in figure 6 (top graph) was chosen by monitoring the Vgk for V2b until it reached about 2.5Vp-p, the same value as the DC bias voltage across R5. At that point, the amplifier’s output voltage into 300 ohms is 5V or 83mW, a six-fold improvement over the 13mW maximum for the original Morgan Jones. The harmonic distortion at 83mW is about 1%.
The performance of the optimized amplifier into a 32-ohm load is as remarkable. Again, the output voltage in figure 7 was chosen by monitoring the Vgk for V2b until it reached about 2.5Vp-p. Based on the results of the White follower balance analysis, the maximum output power of the amplifier into 32 ohms is 10mW, a six-fold improvement over the 1.6mW maximum of the original MJ amplifier, although the harmonic distortion at 10mW is also higher: 2.1%.
The graphs in figure 8 compare the Vgk for the output tubes V2a (blue) and V2b (magenta) for the setups in figures 6 and 7 respectively. When the top graph here is contrasted with the top graph in figure 5, the Vgk waveforms in the optimized amplifier when driving a 300-ohm load have achieved virtually perfect balance. The bottom graph shows that when the optimized amp is connected to a 32-ohm load, the Vgk waveforms are not quite as balanced (the amplitude of Vgk for V2a is slightly larger than that for V2b), but are vastly more balanced than the curves in figure 5.
The frequency responses of the original (top graph) and optimized amplifiers (bottom graph) driving a 32-ohm load are shown in figure 9. The low frequency response of the optimized version has a more extended low-end than the original. The overall gain of the optimized amplifier is about 8, whereas the original has a gain of 19 (the graph scales do not make the differences in gain obvious, however). With the 300-ohm load, the difference in gains is far less: 23 and 19 for the original and optimized amps respectively (graphs not shown). The drop in gain with the 32-ohm load is due to the higher output impedance of the optimized amplifier, one of the tradeoffs of optimization. The original has an output impedance of about 10 ohms, but in the optimized version, the output impedance is 53 ohms – high enough to cause strong loading effects on a 32-ohm load. More than half of the amp’s output voltage is absorbed by the output impedance in this case.
4. The Optimized Morgan Jones Amplifier with Feedback
While the optimized amplifier can dump 10mW into 32 ohms, adding feedback improves the performance even more. The schematic in figure 10 is identical to the one in figure 4, except for the resistors R8 and R9 that form a feedback loop and the removal of the 100-ohm grid stop resistor on V1. A separate grid-stop resistor is not needed, because of the presence of R9. Cavalli selected the feedback resistor values by examining trade-offs. Too much feedback and the gain was reduced too much. Too little feedback and there was no benefit. He tried to select values that reduced the output impedance down substantially below 32 ohms and reduced distortion, while still leaving enough gain to be sensitive to conventional CD inputs. Cavalli recommends experimenting with these resistor values to get the best tradeoff for specific headphones.
The output graphs for this amplifier driving a 300-ohm load are not shown here, but they indicate that it delivers the same power (83mW) as before, but at a lower distortion: 0.5%. Feedback also lowers the output impedance to about 20 ohms and the gain to about 6 for a 300-ohm.
The output characteristics of the optimized MJ amplifier with feedback into 32 ohms are shown in figure 11. The amp’s voltage gain is about 4, because the output impedance, while less, is still significant compared to 32 ohms. Again, the maximum output power under a White follower balance analysis is the same as for the non-feedback version in figure 4, but the distortion is lower: 1.4%. Thus, the primary effect of feedback in this circuit is to provide cleaner output power for low and high impedance headphones.
5. Revised Power Supplies
The power supply used by Johannes Chiu (figure 1) was “bare bones” and provided modest filtering. A 19VAC wallwart directly powered the tube filaments connected in series. A step-up transformer converted the 19VAC to about 156VAC, which was then rectified and filtered with a small 220uF capacitor to output 220VDC.
Bryan Ngiam and Rudy van Stratum have modified the Chiu design to reduce noise and hum. Ngiam built two power supplies (figure 12). The first supply is the closest to the original. It uses an 18VAC wallwart to power the tube filaments connected in series, a different step-up transformer and a L-C pi output filter for improved noise filtering.
Ngiam’s second supply has a single custom power transformer with three 6.3VAC, 500mA secondaries. Like the first supply, the high-voltage primary employs a pi filter, but each 6.3VAC secondary powers the heater of one tube. Ngiam recommends twisting the filament supply wires “to reduce AC currents into the audio circuit.” With either supply, Bryan recommends removing the 1 Megohm and 100 ohm resistors at the input stage.
Rudy van Stratum’s designs are shown in figures 13 and 14, and incorporate power transformers that he already had in his possession: a 250VAC/50mA unit and a 6.3VAC/2A unit. In both circuits, the tube heaters are connected in parallel across the DC filament supply. Like the Ngiam supplies, the high voltage output of Stratum’s first supply uses an L-C pi filter.
For DIYers who cannot find an appropriate inductor, Stratum’s second circuit has several stages of RC filtering with two high voltage outputs: 230VDC and 210VDC. Originally, he powered the entire amplifier off the 230VDC tap. Later, he decided to increase the filtering to the input stage supply by adding a 2.2K resistor and a 100uF electrolytic capacitor, because most of the hum was coming from the input stage. The extra RC filtering also reduced the voltage to about 210VDC. The necessity of two high voltage taps can be avoided if the power transformer is replaced with one that outputs enough voltage to give 220VDC. It might also be worth trying to power the entire amplifier with 210VDC.
Johannes Chiu constructed the original Morgan Jones amplifier. He made one change: a 100-ohm grid stopper resistor to the input tube for each channel to improve stability. The EarMax uses a 19VAC, 350mA wallwart. Chiu created the simple power supply in figure 1 based on a 19VAC wallwart. The 19VAC directly powers the tube filaments connected in series, and could be rectified to DC for lowest noise. To step the 19VAC up to 220V, Chiu used a 10V filament transformer in reverse. However, Frank Nikolajsen pointed out that a 10V filament transformer would actually give a rectified voltage of 310V. Therefore, I have scaled the transformer secondary to 14V, but recommend experimentation. I have increased the wallwart’s current spec to 1A (Chiu’s wallwart had a capacity of 840mA). DIYers in a country with an AC standard different from the US standard should select the transformer accordingly. I have drawn the power supply with a small value filter capacitor (there were no instructions from Chiu about this). DIYers will probably want to increase the value.
Several DIYers have found that Chiu’s supply can introduce excessive hum in the amp. Bryan Ngiam and Rudy van Stratum designed powers supplies with superior filtering (figures 12-14). Bryan recommends that all input cables should be properly shielded and that a star ground should be employed whenever possible. If using a ground “strip” inside the chassis, Stratum suggests experimenting with the positioning of audio grounds on the strip to get the lowest hum and noise.
Alex Cavalli recommends a minimum power supply current rating of 220V @ 50mA for the optimized and feedback amplifiers, figuring 20mA peak per output section and 8mA for the input stages for about 48mA total. Otherwise there will be major power supply sag. He also recommends that the tubes be actual 6922/6JD8. The 6N1P is sometimes subsituted by vendors and is NOT a true substitute (see Bruce Bender’s 6N1P OTL headphone amplifier for a modification of the Morgan Jones design using that tube).
Chiu used a trapezoidal-shaped chassis measuring 2″ (top) x 4″ (bottom) x 6.5″ (length) x 2″ (height), which gives a volume 1.5 times larger than the chassis for the EarMax (3.75″ x 3.5″ x 4.0″). The additional space is required by the filament transformer. The EarMax probably has a smaller custom transformer. For the volume control, Chiu selected a 100K dual audio pot from Radio Shack (RS 271-1732).
The original Morgan Jones amplifier does not have enough current drive for low impedance headphones like the Grados, and Chiu did not have the schematics for the optimized versions. Instead, he experimented with a small impedance matching transformer (1.5K/60-60 from Antique Electronic Supply) for higher power transfer to his Grados. He attached the 1.5K primary to the output of the amplifier and connected the dual secondaries in parallel, which then became the output for the headphones. On the matter of selecting the impedance transformer, Chiu writes:
- People think that the output transformer is crucial in the sound, and hence must be high quality and expensive. HOWEVER, the power involved here to drive the headphones is in the milliwatts. Given that the power is so low, it greatly relaxes the requirements for the transformer. The transformer I got was about the size of a nail, and looks as cheap as many transformers found on computer modems and the like. I would also argue that the specified frequency response is perhaps 100Hz-15kHz, which most people would frown at. BUT, these specs are at full power, which could be up to 1/2 watt. At 10mW, who knows what the response is. All I know is that I tried it, and one would be hard pressed to distinguish and sonic feature that was added because of the transformer. I would encourage people to try out a few transformers. Make sure the impedances are more less correct, such that you have enough current drive, while at the same time not lose too much signal level due to the step down.
Chiu compared his original Morgan Jones amplifier to a LT1010 buffer headphone driver and a tube headphone amplifier (a “monster” 6AS7G cathode follower driven by a 6DJ8 diff amp) he had built earlier. The headphones were a pair of Sennheiser HD420. He found that possibly “because of the higher gain of this circuit, it feels to have more punch and wallop…like moving from a cathode follower pre-amp to a mu-follower, or like adding a huge cap to your pre-amp power supply.”
With a borrowed pair of Grado SR60s and the impedance converter, Chiu noted “I think the transformer will give you 90% of what is there. Maybe the bass is a tad weaker, the highs are different, but still better than straight out from those personal walkman or cd players.” He tried the Grados without the impedance converter “and the transformer is way better.” Chiu’s final verdict on this project: “there is one thing I am certain nobody would deny if they listened to it, and that is: the amp is really fun to listen to.”
DIYers building the Morgan Jones amplifier today should try the optimized versions, which garner favor through lower distortion, higher output power and more stable output stages that benefit both high and low impedance headphones. Although low impedance headphones were starved for current with the original Morgan Jones design, the optimized amps can drive these types of headphones to reasonable volumes without the need for an impedance-matching transformer. Low impedance headphones not only get more power from the optimized amps, but also get a flatter, more extended low frequency response. The mystery of the EarMax Pro, at last, is solved.
Appendix: Simulating the Amplifier in OrCAD PSpice
This section discusses how to use OrCAD Lite circuit simulation software to simulate Alex Cavalli’s optimized Morgan Jones amplifier. OrCAD Lite is free and the CD can be ordered from Cadence Systems. At the time of this writing, OrCAD Lite 9.2 is the latest version. OrCAD Lite 9.1 can be downloaded from the Cadence website (a very large download at over 20M) and should work as well. There are 4 programs in OrCAD suite: Capture, Capture CIS, PSpice and Layout. The minimum installation to run the amplifier simulations is Capture (the schematic drawing program) and PSpice (the circuit simulation program).
Download Simulation Files for Alex Cavalli’s Optimized Morgan Jones Amplifier
Download OrCAD Triode Simulation Libraries
After downloading mj_sim.zip and orcad_triodes.zip, create a project directory and unzip the contents of the mj_sim.zip archive into that directory. Then extract the contents of the orcad_triodes.zip archive into the \OrcadLite\Capture\Library\PSpice directory. The files triode.olb and triode.lib are libraries containing simulation models for several popular types of triode vacuum tubes, including the ones used in this amplifier. They are based on tube SPICE models found at Norman Koren’s Vacuum Tube Audio Page and Duncan’s Amp Pages. Note: heater connections are not required for any of the triode models.
The two basic types of simulation included are frequency response (AC sweep) and time domain. The time domain analysis shows the shape of the output waveform and can be used to determine the amplifier’s harmonic distortion. They both run from the same schematic, but the input sources are different. For the frequency response simulation, the audio input is a VAC (AC voltage source). The time domain simulation requires a VSIN (sine wave generator) input. Before running a simulation, make sure that the correct AC source is connected to the amp’s input on the schematic.
The following instructions for using the simulation files are not a complete tutorial for OrCAD. The OrCAD HELP files and online manuals include tutorials for those who want to learn more about OrCAD.
Frequency Response (AC Sweep) Analysis
- Run OrCAD Capture and open the project file “Morgan Jones.opj”.
- In the Project Manager window, expand the “PSPICE Resources|Simulation Profiles” folder. Right click on “Schematic1-freq_resp” and select “Make Active.”
- In the Project Manager window, expand the “Design Resources|.\morgan jone.dsn|SCHEMATIC1” folder and double click on “PAGE1”.
- On the schematic, make sure that the input of the amp is connected to the V3 AC voltage source. If it is connected to V2, drag the connection to V3. By default, V3 is set to 0.5V. (Note: the tubes in the OrCAD schematic are labelled U1, U2 and U3. In the article schematics, they are referred to as V1, V2a and V2b.)
- To add the triode library to the Capture: click the Place Part toolbar button (). The Place Part dialog appears. Click the Add Library button. Navigate to the triode.olb file and click Open. Make sure that the analog.olb and source.olb libraries are also listed in the dialog. Click the Cancel button to close the Place Part dialog.
- From the menu, select PSpice|Edit Simulation Profile. The Simulation Settings dialog appears. The settings should be as follows:
- Analysis Type: AC Sweep/Noise
- AC Sweep Type: Logarithmic (Decade), Start Freq = 10, End Freq = 100K, Points/Decade = 100
- To add the triode library to PSpice: Click the “Libraries” tab. Click the Browse button and navigate to the the triode.lib file. Click the Add To Design button. If the nom.lib file is not already listed in the dialog list, add it now. Then close the Simulation Settings dialog.
- To display the input and output frequency responses on a single graph, voltage probes must be placed on the input and output points of the schematic. Click the Voltage/Level Marker () on the toolbar and place a marker at the junction of R9 and the grid of U1. Place another marker just above RLoad at the amp’s output.
- To run the frequency response simulation, click the Run PSpice button on the toolbar (). When the simulation finishes, the PSpice graphing window appears. The input and output curves should be in different colors with a key at the bottom of the graph.
- The PSpice simulation has computed the bias voltages and currents in the circuit. To see the bias voltages displayed on the schematic, press the Enable Bias Voltage Display toolbar button (). To see the bias currents displayed on the schematic, press the Enable Bias Current Display toolbar button ().
Time Domain (Transient) Analysis
- On the Capture schematic, make sure that the input of the amp is connected to the V2 sinewave source (the default values are: VAMPL=0.5, Freq. = 1K, VOFF = 0). If it is connected to V3, drag the connection to V2.
- In the Project Manager window, expand the “PSPICE Resources|Simulation Profiles” folder. Right click on “Schematic1-transient” and select “Make Active”
- From the menu, select PSpice|Edit Simulation Profile. The Simulation Settings dialog appears. The settings should be as follows:
- Analysis Type: Time Domain(Transient)
- Transient Options: Run to time = 10ms, Start saving data after = 0ms, Max. step size = 0.001ms
- To display the input and output waveforms on a single graph, voltage probes must be placed on the input and output points of the schematic. Click the Voltage/Level Marker () on the toolbar and place a marker at the junction of R9 and the grid of U1. Place another marker above RLoad at the amp’s output.
- To run the time domain simulation, click the Run PSpice button on the toolbar (). When the simulation finishes, the PSpice graphing window appears. The input and output curves should be in different colors with a key at the bottom of the graph.
- To determine the harmonic distortion at 1KHz (the sine wave frequency), harmonics in the output waveform must be separated out through a Fourier Transform. In the PSpice window, press the FFT toolbar button (). The PSpice graph changes to show the harmonics for the input and output waveforms. The input and output curves should be in different colors with a key at the bottom of the graph.
- The fundamental frequency at 1KHz will have the largest spike. The other harmonics are too small to be seen at the default magnification. In the PSpice window, press the Zoom Area toolbar button () and drag a small rectangle in the lower left corner of the FFT graph. The graph now displays a magnified view of the selected area. Continue zooming in until the harmonic spikes at 2KHz, 3KHz, etc. are visible.
- Harmonic spikes should exist for the output waveform only. The input is an ideal sine wave generator and has no distortion. To calculate total harmonic distortion, add up the spike values (voltages) at frequencies above 1KHz and divide by the voltage at 1KHz (the fundamental).
Additional Simulation Tips
- To change the value of any component on a schematic in the Capture program, double-click on the value and enter a new value at the prompt.
- The schematic included in the simulation files is for the optimized Morgan Jones amplifier with feedback. To simulate the non-feedback amplifiers without removing the feedback resistors, change the values of R8 to 100meg and R9 to 1 ohm (or 100 ohms as the grid-stop Rg).
- To measure the grid-cathode voltage of tubes (Vgk), use the Voltage Differential Marker (). Click the Voltage Differential Marker toolbar button and touch the probe to the tip of the grid pin and then cathode pin.
Note: simulations only approximate the performance of a circuit. The actual performance may vary considerably from the simulation as determined by a number of factors, including the accuracy of the component models, and layout and construction techniques.
c. 2000, 2002 Chu Moy.
10/17/2000: Updated section on power supply.
10/19/2000: Added section on impedance converter.
10/24/2000: Charles King suggested rating the plate resistors at 1W.
4/9/2002: Major revision to article: expanded discussion of original Morgan Jones design and added sections on the optimized Morgan Jones amplifier schematics by Alex Cavelli. Also added section on simulating Morgan Jones circuit using OrCAD Lite.
6/3/2002: Added schematics and descriptions for power supplies by Bryan Ngiam and Rudy van Stratum (figures 12-14). They both built the optimized version of the MJ amp without feedback. Here are pictures of Ngiam’s and Stratum’s amps:
More information about Bryan Ngiam’s power supplies is available at Bryan Ngiam’s Home Page.
3/3/2003: The author updated the 6DJ8 model in the triode library used in the OrCAD simulations of the Morgan Jones amp. He also revised value of R4 from 150 Ohms to 120 Ohms for the optimized amps (figures 5 and 10). He writes:
I’m including a new triode library. I have modified the 6DJ8 model based on some other work that I was doing. The parameters are slightly different from the previous version and they are more accurate. You should use this library and replace it in the ZIP you’re making available on the website. Because of this better model, I am recommending that the plate resistor for the MJ be lowered from 150 ohms to 120 ohms. Closer to 1/Gm.