A Fender-Tone Tube Headphone Amplifier.

by Alex Cavalli

This project grew from several conversations that I had with John Broskie of the TubeCAD Journal. I was looking for a good headphone amp and John and I discussed some of the advantages and disadvantages of several of his published circuits. I must say that for someone who creates so much good material, he is also very generous with his time. One particular discussion led me to a White cathode follower design that seemed elegantly simple, that could easily drive 300ohm headphones, and might do respectable duty driving 32ohm phones. The original circuit appeared in the April-May 2001 issue of TubeCAD Journal.

The more I looked at the circuit, the more I liked it. Naturally, because it was so simple, I decided to make it more complex. An issue for me at the start of the project was that I didn’t have any test and measurement equipment (except my ears), so I needed to find a design that I was pretty sure would work before I committed to building it. I spent many hours simulating various topologies with OrCAD PSpice using triode models from Norman Koren’s Vacuum Tube Audio Page that I modified for better accuracy. I also created a new 5687 model and added it to the library. The more designs I simulated the more I liked the simple White cathode output stage.

While Chu Moy and I were discussing the design for this article, he asked me how closely related Broskie’s amp was to the Morgan Jones amp featured in the HeadWize projects. Thus began a long conversation between us that resulted in the update to the MJ article with the new designs, the OrCAD PSpice circuit simulations and the tube libraries. For Bruce Bender’s 6N1P OTL amplifier, I suggested new values for some of the amplifier resistors and a higher voltage power supply, which Bruce generously agreed to try and was able to get much better results.

The Circuit

The Broskie Headphone Amplifier

Figure 1

Figure 1 is Broskie’s original design for 300-ohm headphones. The plate resistor in this circuit is calculated so that the drive signals to the upper and lower tubes are balanced (see The White Cathode Follower for a discussion about optimizing this configuration). Balancing the drive signals is the biggest issue with this kind of push-pull design. Given the component values, the output impedance of this circuit will be about 53 ohms. It can easily push 20mA peak-to-peak into a 300-ohm load with 0.5V input.

The TubeCAD Journal article at the link above contains a description of this circuit. To summarize here, the output is an optimally designed White cathode follower. A White cathode follower is a push-pull output topology where the signal at the plate of the upper follower is fed back into the grid of the lower triode causing it to follow out of phase with the upper triode (hence the push-pull). To ensure that the tubes both see the same amplitude drive signal the plate resistor must be the reciprocal of the triode’s Gm. The feedback lowers the output impedance below that achievable with a simple cathode follower.

The input stage is a grounded cathode with an active load. Its voltage gain is mu/2, which for the 12AU7 is about 8.5. Broskie selected 5687s for the output tubes because, for miniature triodes, they have very high maximum ratings and good linearity. Maximum current is 30mA and maximum plate dissipation is 3.75W per section. The high maximum current allows the tubes to be biased at 20mA in class A mode. In the push-pull circuit, each triode can swing 10mA up or down with low distortion, giving a total current swing of 20mA. With a Zo of about 53 ohms, the output section has an open loop voltage gain of about 0.92. Furthermore, voltage gain and output impedance stay constant with dropping load impedance.

By comparison, a straight cathode follower using a single 5687 with the same plate voltage, same bias voltage, and open loop voltage gain of 0.76, has an output impedance of 94 ohms. With a 32ohm load the voltage gain drops to .2 and output impedance drops to 24 ohms. Even with a pair 5687s in parallel, the gain only increases to 0.32. Clearly the White cathode follower is a superior power buffer. But even so, Zo of 53 ohms is still not low enough for 32-ohm headphones. The total forward open loop voltage gain of the amplifier is about 7.8 (.92 x 8.5). There are other features of this design relating to power supply noise that are described in the TubeCAD article.

The New Design

As I was pondering this design, several things happened more or less at the same time:

• Although I wanted a respectable tube headphone amp, I also wanted to experiment with some other ideas.
• I realized that I frequently listened to music with some bass and treble boost. I knew that I would not be happy with a new amp if it didn’t have at least some bass/treble boost.
• I realized that the output was out of phase with the input (this is not really as big a deal as some think, but there was an opportunity to correct it anyway).
• I had some extra 5687s, so that if I paralleled output tubes I could lower the impedance by at least half.
• John Broskie published another article in TubeCAD Journal discussing a variety of possible sonic controls that could be incorporated into a tube amplifier design (The Missing Sonic Controls).

This all led me to augment John’s design by doing the following:

• Adding and additional set of output tubes.
• Adding a Fender tone stack.
• Adding an input stage to align the phase and compensate for the loss incurred by the tone stack.
• Adding stereo blend and spatial controls that were now possible between the input stages of each channel.

Figure 2

Figure 2 is the schematic of the final design including the tone stack but leaving out the volume, balance, and sonic controls. The right half of the circuit is John Broskie’s original White cathode amplifier with a pair of 5687s as output tubes. This lowers the output impedance to about 27 ohms (gain remains the same) and allows me to set the quiescent current at 17mA per section to preserve tube life while still achieving high current swings with lower distortion. I’ve increased the size of the output capacitor to get a better lower frequency response at lower load impedances and added good quality audio caps in parallel with the electrolytics. I also added a fuse after the output capacitors to protect the headphones just in case the electrolytic fails someday.

The 10M resistors (R5 and R13) on the grids of the 12AU7s are not strictly necessary. I included them, however, because I was going to build the amp in stages. I knew that at some point I would want to power it up to measure bias voltages before the volume and tone controls were wired. I needed these resistors to ground the grids and set the proper bias points. I could have just as easily used temporary clip leads to ground the grids when measuring voltages, but adding resistors was not that much extra trouble.

The Fender tone stack is a pure boost tone stack. Unlike the equally well-known Baxandall tone stack, the Fender stack cannot provide bass or treble cut. But, since I never listen with bass or treble cut, the Fender stack offered fewer components and complete capacitor coupling thereby avoiding the use of an additional coupling capacitor from the plate of V1. Like all tone stacks, the Fender tone stack has an insertion loss, in this case approximately 18dB. The “boost” actually comes relative to this loss. In the Fender stack, the mid control is only really active if there is bass and/or treble boost. When the bass/treble controls are off the mid control becomes another volume control.

The parts of the stack form several high-pass filters. The upper part is a high-pass filter through the 250pF capacitor into the treble control whose wiper feeds the next stage. The high-pass bass filter (through the 0.47uF capacitor) sits under the treble control. The high-pass mid filter sits under the bass control. Any signal at the top of the bass or mid controls is fed directly to the treble control’s wiper. If the bass control is all the way on, then the low frequencies are passed directly to the wiper. If the bass control is all the way off (zero resistance), then the low frequencies are shorted through the mid control to ground.

The mid control has a higher pass frequency than the bass control. It bleeds off some of the signal that otherwise would pass through the bass section to ground. Because the turnover point is higher than the bass turnover point, and because, when there is treble boost, high frequencies go through the treble section, the mid control has the effect of creating a mid-range notch that in this design is at about 400Hz-1KHz depending on bass/treble settings.

When bass and treble are off, the output sides of the lower two capacitors and the treble control wiper are shorted together and the response is basically flat. The mid control acts simply as a resistance load which, when set to zero, shorts the entire signal to ground. The 27K resistor prevents V1 from seeing a pure AC short to ground. An extremely useful tool for calculating the characteristics of various tone stacks is the Tone Stack Calculator from Duncan’s Amp Pages. I used this tool when designing this circuit.

Because the Fender stack introduces an 18dB loss, it was necessary to recover some of this loss with the additional input stage. This stage has a voltage gain of about 5.5 (with a bypass capacitor the gain would have been a little over 10). It does add some distortion to the original design (which has very low distortion), but I was willing to accept this to experiment with these additional controls. When bass/treble are off and mid is halfway the total open loop voltage gain is about 7. This is plenty of gain for CD players.

I had to compromise the design here because the spatial control (described below) requires unbypassed cathode resistors. To get enough gain from the first stage I used a fairly high plate resistor. Together these features give the input stage an output impedance of about 8K. The input impedance of the tone stack is about 27K at its worst (everything turned off). A higher ratio of impedances would be better, but I decided not to mess with the design any further than this.

Figure 3

Figure 3 shows the input section with the balance, volume, and the spatial and blend sonic controls. The balance pot has an “M-N” taper, where at center position, there is zero resistance in each channel The output is shorted for half of the pot rotation and is linear taper for the other half, and each section operating in reverse compared to the other channel. Turning the control to the left attenuates the right channel without affecting the left channel and vice-versa. The blend control performs a standard stereo-mono blend between left and right channels by more or less shorting the plates of V1A and V1B together.

The spatial control is quite interesting. It introduces an in-phase signal from one cathode to the other. When an in-phase signal is applied to the cathode of a grounded cathode stage, this is negative feedback. Here’s an example that helps to explain its operation. In the “soundstage,” a source to the left of center will have a larger presence in the left signal than in the right, but will have a presence in both channels (under normal conditions). Feeding an in-phase left signal into cathode of the right channel will cancel some of source signal that is already in the right channel. And vice-versa. This will tend to make the channels sound more separated or “farther apart” because they no longer contain as many common sounds. [Editor: this type of spatial effect is also called ambience enhancement.] I am able to hear this effect, although it can be very subtle.

Power Supply

Figure 4

The power supply, Figure 4, is a simple solid-state supply, although it is slightly overbuilt. During simulation I discovered that if the plate voltage for the input stage were drawn from the B+ (high voltage tap) of the output stage the circuit would oscillate, even with standard isolation techniques (R-C filtering). The prototype amp hummed slightly. I put the 5H choke in to track down the hum problem. In the original power supply, I tapped the first stage B+ from the very first filter section. When I installed the choke I had to take all four filter sections from the output of the choke. Both channels have the first two filter stages in common, but different final filter sections.

The heater supply uses a 12V, 5A low-dropout regulator (Linear Technology LT1084-CT-12). One problem with the amplifier design is the range of heater-to-cathode voltages present. If the heater supply were grounded the upper tubes would have heater-to-cathode voltages greater than 125VDC, exceeding 100V specification for the 5687. I solved this problem by borrowing a technique from Bruce Rozenblit (see the OTL design in his book, Audio Reality) where the heater circuit is allowed to DC float, but is AC grounded through a capacitor at the mid-point of V1’s heater. In my case, I attached the capacitor to the negative rail of the heater supply.

There are two power switches; one to turn on the heater supply and the other to turn on the B+ supply. The B+ switch is wired to the heater switch as a fail-safe. Studying the circuit I can’t see any places where adverse voltages would be applied if the B+ came on before the tubes were conducting. But, I turn the heaters on first anyway.

CONSTRUCTION

Component Sources:

Allied Electronics – chassis
Antique Electronics – knobs, audio caps, RCA jacks
Audio Electronic Supply – Noble balance control
Avel-Lindberg – toroidal power transformers
Digikey – wire, regulators
Mouser – resistors, capacitors
Radio Shack – switches, fuses, and assorted parts
TubeBuilder – terminal boards
TubeWorld – matched 12AU7s and 5687s

Stage 1

The front of the amplifier is shown at the beginning of this article. The input and output jacks are on the far left, the power switches on the right. The inputs include right and left RCA jacks and a single stereo mini-jack wired together appropriately. The outputs are both a 1/4″ phone jack and another stereo mini-jack also wired together appropriately. The controls are not labeled yet, but they are from left to right: balance, volume, bass, treble, mid, blend, spatial.

First, I constructed both power supplies and the Tubebuilder boards (ordered from TubeBuilder.com). Laying out a TubeBuilder terminal board is much like laying out a PC board, except the runs are made with point-to-point wiring and the components are connected directly to the tube sockets or the binding posts. Each board has tube sockets, binding posts, and copper ground plane with standoff mounting spacers underneath. I really like the Tubebuilder terminal boards because they let me do a lot of the wiring outside of the confines of the chassis. It’s much easier to wire the boards first, install them, and complete the power and signal wiring.

Then I drilled, punched, and painted the chassis. Unfortunately, I experimented with the paint on this project. I used appliance-grade black paint whose durability is not what I had hoped for. If I were starting again, I would use standard spray paints and good quality primers that can be found at most hardware stores. Since the chassis is metal, I would bake the paint on carefully. The aluminum chassis and cover plate are made by Hammond. The chassis dimensions are 17″ x 3″ x 10″. Once I painted the chassis black, I decided to paint the screws blue. Since the transformers and caps were blue, it was an easy choice for the screw color.

In the first prototype, there was a small fan on the top of the chassis to provide ventilation. I’ve removed the fan completely because it was too noisy. Now that the regulator is heatsunk to the chassis, the ventilation is not really necessary. Since I had a big hole in the top of the chassis, I made a cover plate with holes in it to let some of the internal heat out anyway. It doesn’t really have much effect, but I had to cover the hole somehow.

The sockets on the Tubebuilder boards are connected to the copper ground plane, which I grounded to the circuit ground. Thus, it is good practice that the tube sockets not touch the chassis to avoid ground currents. Normally the Tubebuilder boards are mounted so far beneath the surface (see below), with only the tubes sticking out, that this is not a problem. But I wanted to be able to see all of the tubes, so I shortened the spacers making it possible for the sockets to electrically contact the chassis. To fix this I drilled an extra large hole for the input tube and cut rectangular openings for the power sections using a saber saw. I filed the hole and cutouts to clean them up. Then I drilled all of the rest of the mounting holes and painted.

The toroidal power transformers were from Avel-Lindberg. (A-L were extremely helpful in taking their time to help me with a very small order of just two transformers.) They are mounted on the top of the chassis, because I like the way they look and because the heater transformer gets very hot. Tube circuits are already hot. No sense subjecting the internal enclosed components to more heat than necessary. Wires for the transformers are fed through rubber grommets.

The heater rectifier and regulator are heat-sunk to the inside of the chassis. Between the two of them they dissipate a lot of power and after a while the entire chassis is pretty warm. It might be possible to use a 3A regulator in place of a 5A regulator, because the heater requirements are 2.7A. However, the “cold heater” current of the tubes may cause the initial surge current to exceed 3A. Current limiting regulators will shutdown and may prevent any heater current from flowing to warm up the tubes so that the current demand can then drop below 3A.

However, in either case, the regulator must be a low-dropout type regulator. Standard regulators must have an input voltage at least 2.5V above the output voltage. But, unless you’ve got a huge 12V transformer, the voltage input to the regulator is likely to drop to or below 14.5V. At this point a standard regulator will stop regulating. On the other hand, a low-dropout regulators need only about 1V difference input to output. In my case, under load, the input to the regulator is about 14.4V, on the edge for a normal regulator, but well within specs for a low-dropout type.

The tone stack capacitors are inserted into the perf board that is suspended in mid air near the controls on the front. The input stage Tubebuilder board is underneath the perf board. The output stage high voltage supply is on the right and the input stage and heater supplies are on the perf board on the left. The non-electrolytic output capacitors are in the center with the fuses.

An aluminum shield installed between the high voltage supply/transformer and the rest of the circuit reduces radiation problems. After mounting the power supply and Tubebuilder boards to the chassis, I wired up the as much as can be wired without connecting the jacks and controls (this includes wiring the power switches). The power cord ground and the circuit chassis ground are connected to a single chassis ground point.

I tested the 12V supply first without tubes, including testing to make sure that it was floating. I installed all of the tubes and tested the heater supply again and powered up the high voltage and waited for smoke. When none appeared, I started measuring the bias voltages according to figures 2 and 4. In my case, the actual B+ voltages were little higher than designed, but well within tolerance. When these checked out, I pulled the tubes and started Stage 2.

Stage 2

At this point I mounted the jacks and controls, positioning their various connections for ease of wiring. All of the jacks’ grounds are insulated from the chassis using plastic and fiber washers, again to avoid ground loops. The Noble balance pot has a small circuit board that is mounted to its pins. Wiring is then done to the circuit board.

Good design made the control wiring very simple. Once I got this far, I built the tone control board. This board contains the three tone capacitors and the 27K resistor for both channels. The board is mounted in mid-air above the input stage (or below depending on your orientation). Because it is so light, the board is held in place by the wires connecting it to the rest of the circuit. I used shielded wire for the signal wires to an from the tone stack and from the volume control to the input stage.

The last step was to mount and wire the output electrolytics. To do this, I had first drilled holes in chassis about 1/4″ smaller diameter than the capacitor itself. Then, as part of early construction, I mounted a perf board flush underneath the chassis covering these holes. I used the screws that fastened the perf board to also attach the fuse holders. The caps are snap-mount types with fairly short, stiff leads. I pushed the leads through the perf board from the top and bent them over, which essentially fastened the caps down to the chassis. Then I wired the parallel audio caps and connected these pairs to the output of each channel, through the fuses to the output jacks.

Anything resembling a wire bundle is held together with plastic cable ties. Once the controls and tone board were wired, I attached the knobs. The knobs are available from Antique Electronics (see below). They are 1″ black anodized aluminum. I also bought the Solen caps and the gold-plated RCA jacks from these folks.

Except for the balance control, the pots are all inexpensive Alpha potentiometers that I bought from Mouser. The dual balance pot is Noble part number 220Y100K(W)X2-9920, purchased from Audio Electronic Supply. ALPS also makes this kind of pot, but I couldn’t find the ALPS pots on the web, so I bought the Noble. If you had to substitute, you could just use an ordinary dual 100K linear pot wired as a balance control. This will cut the signal strength in half immediately at the input, but there is plenty of gain to compensate for it. I just don’t like to do this because gain always adds distortion and noise.

The values for the “mid-point” resistance of the blend and spatial controls, calculated using the formulas in the TubeCAD Journal, are around 10K. After some experimentation I finaly chose a 50K linear pot for the blend control and a 25K linear pot for the spatial control. Both pots have rear-mounted switches so that the controls can be completely removed from the circuit.

The small capacitors in the amp (figure 2) are metal film, except for C1. C1 is a ceramic capacitor, because I was not able to find a 250pF film capacitor. I have heard the various discussions about capacitors for audio equipment. But, I am not enough of an expert on this to know one way or the other. Plus, I don’t have any equipment and I am not able to try experimenting with various caps. I did later see some 250pF silver mica caps at Antique Electronic Supply and would have used them just to be safe. In truth, however, I don’t know how much difference it would make. The amp sounds OK to me and the highs (where these caps matter) don’t sound harsh or brittle or whatever.

Here are the component layout and wiring diagrams for the TubeBuilder boards:

PERFORMANCE

As I mentioned above, I don’t have test and measurement equipment so my specs on this circuit are derived from simulations (which are generally better than reality). The performance of the amp, as determined from simulation, is shown in figures 5 through 8.

Note: the gain of the amplifier, described above, is not really that important. After all if the 0.5 volt output from a CD player line-out could drive 32-ohm headphones with enough current, it would be delivering 8mW (pretty loud). What any power amp really does is convert the voltage at the input into current into the load. As everyone knows power amps are just buffers that translate a high-impedance voltage source into a low impedance current source. In figures 5 to 8, therefore, I’ll be looking at current into the load.

Figure 5

Figure 5 shows the current supplied to a 300ohm resistive load. The input is 0.5V. The bass and treble controls are set at midpoint (in terms of resistance, not rotation). The midrange control is all the way on. The power output into the load (using P = (I^2 * R)/ 2) is about 15mW. For most headphones this should be more than adequate power.

Figure 6

Figure 6 shows a Fourier analysis of this same trace. I’m using the Fourier analysis to estimate the distortion. For those who don’t know what a Fourier analysis is, a Fourier transform translates a time-varying signal into the frequency domain showing all of the frequency components that comprise the signal with their respective amplitudes. In this case, the input signal is a pure sine wave at 1KHz. If any other frequency components show up in the current at the load, then the amp is adding harmonic distortion (harmonic because the frequencies occur at harmonics of the base signal).

Before reviewing the Fourier analysis, it is important to point out that this is simulation. Real circuits exhibit many characteristics that are not accounted for in some of the models, including the fact that simulated components are perfect unless specifically created to be otherwise. In this case, the resistor and capacitor models are perfect; the distortion is coming primarily from the non-linearities in the tube characteristics. We should use these estimates with caution, although they should give a basic picture of what the amp is doing.

The large peak at 1KHz is, obviously, the primary signal (it goes up off the screen). There is no significant signal at the fourth harmonic or above. The total current flowing at 2KHz and 3KHz (the small peaks) is about 3ua. An estimate of the THD is given by dividing this number by the total current flowing at 1KHz, about 10mA. Thus, 3uA/10mA = 0.0003 = 0.03%. This is a very low distortion figure. The real number will be higher than this. The only way to get this number is to measure it. However, the simulation indicates that the amp should have very good distortion figures. Even if the real distortion is ten times higher, it will be around 0.3%. Still a good number.

Figure 7

Figure 7 shows the current supplied to a 32-ohm resistive load, 0.2V input at 1KHz. The bass and treble controls are set at midpoint. The midrange control is all the way on. At 22mA into 32 ohms, the amp is producing 8mW. This is still a lot of power for most headphones.

Figure 8

Figure 8 is the Fourier analysis for this output trace. Adding up second and third harmonics we have about 85ua. Performing the same calculation as before, the THD is approximately 0.38%. The actual distortion may be higher than this. As you can see the amp does better with a 300-ohm load than a 32, but it still performs well at 32 ohms.

At 15mW and 8mW power delivery into the loads of Figures 5 to 8, the amp does not exhibit any clipping behavior. This is because the quiescent current in the output section is about 36mA, making it possible to swing ±32ma in class A operation with minimal distortion and without clipping. Beyond this, the grids in the real circuit will start to be driven positive. The simulation models do not show the effect of this very accurately.

Frequency Response

Figure 9

Figure 7 shows the amp driving a 32-ohm load at 40mA peak-to-peak. At 20mA peak-to-peak, THD is about 0.15%. The THD in both cases seems fairly constant from 20Hz-20KHz. With the tone controls set at no boost, response is better than 3db flat from 10Hz to 300KHz, shown in Figure 9 (PSpice AC analysis result).

Figure 10

Figure 10 is the frequency response with bass and treble at full boost and the mid control at half. The volume control for this simulation was set lower to keep figures 9 and 10 near the same scale. The center of the notch is about 400Hz.

RESULTS

My current headphones are Sony MDR-V600 Studio Headphones (45 ohms) that can be found at most consumer electronics outlets. My music source is a Sony portable CD player line output. There is much to be desired in replacing this combination, but even with these components I can say that the sound from the amplifier is very clean and crisp. It is also very quiet. The tone controls give me enough flexibility to satisfy my listening habits. The stereo blend control works exactly as expected. At first, I couldn’t really hear the effect of the spatial control. But after listening carefully, I can hear the sound stage get wider and narrower as the control is brought through its “mid-point” value. The amount of effect, of course, depends on how the music was recorded to begin with.

The performance numbers from simulation are borne out to the extent that with 45-ohm headphones I can barely crack the volume control to achieve a comfortable listening level, even with the tone stack set flat. With the volume near halfway, the sound level is unbearable for me. This leads me to believe that this amp could adequately drive Grado 32-ohm headphones. I have tested the cheap 32-ohm headphones that come with portable CD players. The amp easily drives these.

If I were to build another amp like this one, I would add a crossfeed section based on the designs described in HeadWize, and I might experiment with some other tone controls and equalizer designs. Actually, I am very happy with the Fender tone stack. I have found a combination of bass/treble boost that I like (as I thought I would) with my Sony headphones: in terms of rotation – Bass 1/2, Treble 3/4, Mid 1/2. This is a very pleasing sound for me and I find myself leaving this combination for almost all recordings. The effect of the spatial control is to separate the soundstage. It is not that substantial. Furthermore, I generally don’t have problems with headphone ambience. I guess I’m not a true headphone connoisseur!

I would probably use the Fender stack again. I would eliminate the ambience control and design a crossfeed filter. I might also remove the regulator on the heaters and just put a giant electrolytic (10,000 – 22,000uF). This would be easier, although it would require a voltage dropping resistor that would still dissipate a lot of power.

The next step for me is to get some better headphones and music source! In fact, if someone wants to contribute an article on building a tube CD player, “I’m all ears!” In the meantime, this project has exceeded all of my expectations. My thanks again to John Broskie for his great work in the TubeCAD Journal and for taking time to give me his insightful comments.

Appendix: Simulating the Amplifier in OrCAD PSpice

Alex Cavalli has provided the project files for simulating this amplifier using OrCAD Lite circuit simulation software. OrCAD Lite is free and the CD can be ordered from Cadence Systems. At the time of this writing, OrCAD Lite 9.2 is the latest version. OrCAD Lite 9.1 can be downloaded from the Cadence website (a very large download at over 20M) and should work as well. There are 4 programs in OrCAD suite: Capture, Capture CIS, PSpice and Layout. The minimum installation to run the amplifier simulations is Capture (the schematic drawing program) and PSpice (the circuit simulation program).

Download Simulation Files for Cavalli Headphone Amplifier

Download OrCAD Triode Simulation Libraries

After downloading cavalli_sim.zip and orcad_triodes.zip, create a project directory and unzip the contents of the mj_sim.zip archive into that directory. Then extract the contents of the orcad_triodes.zip archive into the \OrcadLite\Capture\Library\PSpice directory. The files triode.olb and triode.lib libraries contain simulation models for several popular types of triode vacuum tubes including the ones used in this amplifier. They are based on tube SPICE models found at Norman Koren’s Vacuum Tube Audio Page and Duncan’s Amp Pages. Put the files triode.olb and triode.lib into the \OrcadLite\Capture\Library\PSpice directory. Note: heater connections are not required for any of the triode models.

The two basic types of simulation included are frequency response (AC sweep) and time domain. The time domain analysis shows the shape of the output waveform and can be used to determine the amplifier’s harmonic distortion. They both run from the same schematic, but the input sources are different. For the frequency response simulation, the audio input is a VAC (AC voltage source). The time domain simulation requires a VSIN (sine wave generator) input. Before running a simulation, make sure that the correct AC source is connected to the amp’s input on the schematic.

The following instructions for using the simulation files are not a complete tutorial for OrCAD. The OrCAD HELP files and online manuals include tutorials for those who want to learn more about OrCAD.

Frequency Response (AC Sweep) Analysis

1. Run OrCAD Capture and open the project file cavalli.opj.
2. In the Project Manager window, expand the “PSPICE Resources|Simulation Profiles” folder. Right click on “Schematic1-Response” and select “Make Active.”
3. In the Project Manager window, expand the “Design Resources|.\cavalli.dsn|SCHEMATIC1” folder and double click on “PAGE1”.
4. On the schematic, make sure that the input of the amp is connected to the V4 AC voltage source. If it is connected to V3, drag the connection to V4.
5. To add the triode library to the Capture: click the Place Part toolbar button (). The Place Part dialog appears. Click the Add Library button. Navigate to the triode.olb file and click Open. Make sure that the analog.olb and source.olb libraries are also listed in the dialog. Click the Cancel button to close the Place Part dialog.
6. From the menu, select PSpice|Edit Simulation Profile. The Simulation Settings dialog appears. The settings should be as follows:
   Analysis Type: AC Sweep/Noise
   AC Sweep Type: Logarithmic (Decade), Start Freq = 10, End Freq = 300K, Points/Decade = 100
7. To add the triode library to PSpice: Click the “Libraries” tab. Click the Browse button and navigate to the the triode.lib file. Click the Add To Design button. If the nom.lib file is not already listed in the dialog list, add it now. Then close the Simulation Settings dialog.
8. To display the input and output frequency responses on a single graph, voltage probes must be placed on the input and output points of the schematic. Click the Voltage/Level Marker () on the toolbar and place a marker at the junction of R6 and the grid of U7. Place another marker above R_L at the amp’s output.
   
9. The tone controls are set full on, so the frequency response is not flat. To get a flat response set R10A to 250K, R10B to 1, R11 to 1, and R12 to 5K. The treble pot has to be represented by two resistors because PSpice doens’t have a native variable resistor model.
10. To run the frequency response simulation, click the Run PSpice button on the toolbar (). When the simulation finishes, the PSpice graphing window appears. The input and output curves should be in different colors with a key at the bottom of the graph.
11. The PSpice simulation has computed the bias voltages and currents in the circuit. To see the bias voltages displayed on the schematic, press the Enable Bias Voltage Display toolbar button (). To see the bias currents displayed on the schematic, press the Enable Bias Current Display toolbar button ().

Time Domain (Transient) Analysis

1. On the Capture schematic, make sure that the input of the amp is connected to the V4 sinewave source (VAMPL=0.25, Freq. = 1K, VOFF = 0). If it is connected to V3, drag the connection to V4.
2. In the Project Manager window, expand the “PSPICE Resources|Simulation Profiles” folder. Right click on “Schematic1-Transient” and select “Make Active”
3. From the menu, select PSpice|Edit Simulation Profile. The Simulation Settings dialog appears. The settings should be as follows:
   • Analysis Type: Time Domain(Transient)
   Transient Options: Run to time = 80ms, Start saving data after = 40ms, Max. step size = 0.001ms
4. To display the input and output waveforms on a single graph, voltage probes must be placed on the input and output points of the schematic. Click the Voltage/Level Marker () on the toolbar and place a marker at the junction of R6 and the grid of U7. Place another marker above R_L at the amp’s output.
5. To run the time domain simulation, click the Run PSpice button on the toolbar (). When the simulation finishes, the PSpice graphing window appears. The input and output curves should be in different colors with a key at the bottom of the graph.
6. To determine the harmonic distortion at 1KHz (the sine wave frequency), harmonics in the output waveform must be separated out through a Fourier Transform. In the PSpice window, press the FFT toolbar button (). The PSpice graph changes to show the harmonics for the input and output waveforms. The input and output curves should be in different colors with a key at the bottom of the graph.
7. The fundamental frequency at 1KHz will have the largest spike. The other harmonics are too small to be seen at the default magnification. In the PSpice window, press the Zoom Area toolbar button () and drag a small rectangle in the lower left corner of the FFT graph. The graph now displays a magnified view of the selected area. Continue zooming in until the harmonic spikes at 2KHz, 3KHz, etc. are visible.
8. Harmonic spikes should exist for the output waveform only. The input is an ideal sine wave generator and has no distortion. To calculate total harmonic distortion, add up the spike values (voltages) at frequencies above 1KHz and divide by the voltage at 1KHz (the fundamental).

Note: simulations only approximate the performance of a circuit. The actual performance may vary considerably from the simulation as determined by a number of factors, including the accuracy of the component models, and layout and construction techniques.

c. 2002 Alex Cavalli.

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A DC-Coupled Tube Amplifier With Futterman Output Stage for Dynamic Headphones.

by Rudy van Stratum

[Editor: Although this headphone amplifier design has a direct-coupled output, please take note that the author cautions against using the DC output with low impedance headphones. The potential drifting of bias voltages at the DC output could damage headphones. DIYers who want to avoid this risk should use the AC-coupled output instead.]

I published a beautiful headphone design in 1995 in the Dutch magazine Audio & Techniek (which does not exist any more). It is a Single-Ended Push Pull (SEPP) OTL design a la Futterman, and the whole idea was to get a complete tubed amplifier without an electrolytic capacitor in the signal path. Therefore the design is quite complex, especially in the power supply. The charm of the Futterman is that it is the only design with tubes that can do away with the output cap completely (or it allows the use of non-polar MKP caps or even low voltage electrolytic caps, which can be tried in big quantities for low prices to find out the optimum).

Background

In 1994 I built an electrostatic headphone driver with tubes for my Stax and Micro-Seiki MX-5 headphones. I used the design of Joseph Curcio as published in Glass Audio (volume 1, number 0, 1988). The sounds were fabulous and the effect was that I did not listen anymore to normal dynamic headphones.

Then I got the idea to build a headphone driver for dynamic headphones. Only then could I compare the electrostatic and dynamic types on a more honest basis. I wanted to use tubes as I’m ‘into’ tubes for a very long time (started with an old Quad II very long ago). The Curcio also used tubes throughout and did not need an output transformer. So the design criteria were: tubes, no output transformer (OTL), simplicity.

At that time I had just built an OTL amplifier for normal 8 Ohm speakers using only 3 tubes (EF86, ECC81 and one 6336A double-triode, published in Dutch magazine Audio & Techniek 1994). I had studied a lot of old material on the OTL subject. I decided to go for a ‘standard’ Futterman design using two ECC88’s per channel. And I thought at the time that I did not want an electrolytic capacitor at the output (you always can hear such a device, so when possible avoid it). Just when I was building the prototype there was an article on a schematic for dynamic headphones in Glass Audio. The article from Denzil Denner (Glass Audio volume 6, number 4, 1994) used one ECC83 and one 12BH7 and looked a bit like what I wanted to do. It used a Futterman-like SEPP output stage but had its own way to overcome the problems associated with it. There was an electrolytic cap at the output and I think with the 12BH7 it was optimised for use with mainly 600 Ohms headphones.

The Circuits

FIGURE 1

The schematic of my Futterman headphone amp design is shown in figure 1. A common-as-muck anode-follower amplifies the input signal (gain = 25) and the other half of the first ECC88 is a cathodyne phase splitter using relatively low resistor values of 22k. There is a lot of literature on the patent that Futterman got in 1954 about how to optimally feed the SEPP stage. This stage needs two out-of-phase inputs at the two grids. Because one half of the SEPP does amplify and the other does not, now the problem is how to get the two inputs of the right but unequal amplitudes. Denzil Denner in his Glass-1994-circuit solves this problem by using a simple voltage-divider for the lower half of the SEPP-stage. This will do and is the most simple solution but it will work optimally only for certain loads (which will vary anyhow). And what is more, Danner uses a feedback setting in his circuit that implies the use of an extra electrolytic at the input of the circuit (one electrolytic is bad enough). We must stay open minded however, I did not build the Denner-circuit and can not have a judgment on the sound quality it delivers (and to be honest: maybe electrolytics and output transformers are not that bad after all, but that is for another story).

Returning to the problem of how to drive the SEPP, the genius-like solution of Futterman was to ground the phase-splitter via the loudspeaker to ground. In the schematic you see the 22k resistor in the cathode of the phase-splitter go to the output point of the SEPP stage. The output is also fed back to the input tube by the 1 kOhm resistor.By the way, Futterman circuits are renown of their heavy use of negative feedback. In the original articles of the fifties Futterman is very proud of using 50 dB or more of negative feedback without the amplifier going into some form of oscillation. I must confess that the secrets of the workings of the Futterman-patent are not completely transparent to me yet, there still lacks a modern and clear explanation of it (have seen most of the literature on it, but .. ).

FIGURE 2

So far so good. Very simple. The power supply is the price you have to pay. Because I wanted DC operation at the output I need a symmetrical power suppy for the SEPP stage. So I have plus and minus 90 Volts to start with. Next I want to fiddle with the quiescent currents of the SEPP stage to regulate the DC offset. I use four separate windings of 4 Volt that after rectification allow me to regulate the bias of four tube-halves separately. Further, there is the normal high voltage supply for the first tube and an arrangement for the heaters (12 Volt regulated by a 7812 in a parallel series set-up).

Construction

I used Philips E88CC SQ vacuum tubes (special quality gold pin). Other good brands include Siemens, Telefunken, some Russians like Sovtek, Chinese Golden Dragon etc. There are a lot of good tubes around; I prefer New Old Stock of European brands.

Matching the output tubes is not necessary, because the 4 separate pots can correct a lot of mismatch (which of course does not benefit the maximum output power and THD). Matching is best done by exchanging tubes and looking at the voltages at different points in the circuit. But, most of the time, matching is commercial marketing to sell more expensive tubes. I tried more than 10 tubes in the circuit and they all worked all right. Differences between halves within a tube are not worth the expense and trouble in this circuit. It is better to use 4 tubes from the same same brand/year (or 2 of the same for left/right input and 2 of the same for left/right output). The check is whether the DC output voltage is around 0 volts. If not, then swap the tubes from one place to another and try again. Note: matching the output tubes will not prevent DC drift.

The resistors are all standard Philips metal film 1 Watt, except where otherwise noted. The bias adjustment pots vary the currents of upper and lower half of the SEPP output stage (the 2K pots in the power supply). They are CRUCIAL. In effect, they are operating in a floating sense and are not attached to ground anywhere. You can use any pot, but I used Bourns precision pots that work just fine with their 10 rotations.

All electrolytic capacitors are from Philips (blue colored). The signal caps (0.47uF) are ERO MKT, 630 Volts. The output cap (for the ‘safe’connection) can be either a 100uF, 100V MKP or MKT capacitor (the very big one that is black colored in the photo is an Audyn 100uF MKP) or a high quality 330uF/16V electrolytic. MKT capacitors have a polyester dielectricum; MKP uses a polypropylene dielectricum. MKP caps are about twice the price of MKT. Nevertheless, it depends on brand and make which of the two sounds better; everyone has to judge for himself. More than once, I had MKT caps that did sound better than the more expensive MKPs. For electrolytic output caps, I mostly use Philips electrolytic caps of 16V, 25V or 35V types. These are very fine for the money (blue tubular caps).

Figure 3

I ordered a custom-made power transformer with 7 windings. Or several transformers together can do the job. For example, you can use the combination of a standard tube transformer for the high voltage and filaments, a toroidial transformer (70-0-70 VAC) for the bipolar supply and two cheap transformers that give 4 VAC for the bias section (such as those used for toy trains and the like). Maybe you can even use batteries to supply the bias voltages. I think you can use a simpler bias arrangement for both channels (figure 3) that uses just two 4VAC secondaries, but I simply don’t know. I can only assure the good workings of figure 2, because that’s is the one that I used for many years.

To rectify the ±90VDC supply and the 220VDC high voltage supply, I used 1N4007 diodes. To rectify the bias set-up voltages, I used small bridges of type B40C800 (total of 4 pieces for 2 channels). For the heater supply, I used a bridge rectifier of 10 Amperes that I fixed to the bottom of the chassis (B40/10 and the like).

I used a cooling plate for the 7812 voltage regulator (TO3 version) that I mounted to the chassis. I used a general power supply fuse, around 500mA for 230VAC or 1A for 120VAC. I think this fuse is too slow to have any effect if something is going wrong (it can handle 110 Watts!). I always take a fair margin and take slow fuses; this is not the safest way, however. For safety, one can take a fast fuse and probably 250-500 mA will do (just try and see; if the fuse stays in, it is okay). The on-off switch is a standard toggle switch.

I did not encounter any instability during the construction process. As seen in the photo, I started with the power transformers, then a separate euro-card with all power supply parts (in a logical order) and then another euro-card for the amplifier circuits. I kept the signal wires as short as possible and hard-wired all components to each other using the back-side of the Euro-card. For wiring I used mostly OFC-copper wire.

I made the chassis myself using all kinds of aluminium parts. The dimensions are around 44 x 12 x 10 (cm) (all internally measured). The chassis has a perforated cover not shown in the photos.

Biasing the Output Stage

The first 2 stages are auto biased. So the only variables are bias 1 and 2 of the output stage. Using an ECC88 the optimal bias for the minus pole is around 1.5 Volts leading to 15 mA of current in the lower half of the ecc88. So BIAS 2 = 1.5 Volt. For the plus pole the optimal bias is around 2 Volts leading to a quiescent current of 12 mA for the upper half of the ECC88. The difference in quiescent currents between plus and minus just makes the amplifier putting out 0 mV of offset at the output.

I’ve put in small 4.7 Ohm resistors in the SEPP where you can measure the idle current. Ideally you measure 55 mV and 70 mV over these resistors to get 12 mA for the upper half of the SEPP (that goes to the plus 90 Volt) and 15 mA for the lower half of the SEPP (that goes to the -90 Volt supply). The corresponding biases for the ECC88-halves are around -2 and -1.5 Volts respectively. The best way to go is to set up both biases to these voltages and tune one of the biases till you have 0 mV offset at the output. (Note: why the currents of upper and lower halves are not the same? Because if you try to do this you get several volts on the output, a characteristic of this Futterman solution. In a normal 8-Ohm Futterman amplifier this can not easily be seen because the phase-splitter draws almost no current compared to the big SEPP output tubes of the like of EL519’s).

If you use the output with the cap there is no need to adjust anything at all afterwards. The DC output however drifts away around several tens of millovolts and needs to be used carefully. I mainly used this output to check the sound during short periods, not recommended for using ‘blindly’ at any time.

I did not use a volume pot in the amp itself. I use a separate tube pre-amp to do the job. Adding a volume pot into the amp itself is of course no problem. I would use an Alps 100k dual pot (a blue one, around 25 dollars); these work very fine.

Results

I built this amplifier in 1994 and published the schematic in 1995. It worked the first time and I have listened to it for 7 years very nicely. The sound is very good and you can put in any phones you like (32 Ohms or higher). The headphones shown in the top photo are Yamaha YHD-3 (300 ohms). I used during several years the following ones: Sennheiser HD580 precision (300 Ohms), Sennheiser HD433 (40 Ohms). The best results I got with Sennheiser HD600 and AKG K-240 phones.

As an amateur I have limited capabilities to measure things, but I guess the output impedance of the design is about several ohms due to feedback and the SEPP-arrangement. Because it is a real push-pull topology, it can swing out more than 20 mA easily. I don’t have measuring equipment, but made these calculations of output power:

30 Ohms	13.5 mW in class A, 24 mW in class B
100 Ohms	45 mW/80 mW
400 Ohms	180 mW/320 mW
600 Ohms	270 mW/480 mW

I never had any problems with phones going not loud enough. For the power hungry phones you could think of using the very special E288CC tubes that doubles the capacity of the now used ECC88/6922/6DJ8. The very expensive and scarce E288CC can be plugged into this circuit without modification (you have to set-up the idle currents anew though).

Later I bought the X-cans of Musical Fidelity, just to have something comparable. These X-cans can put out 100 mW at 40 Ohm and I wonder how they do that with only one ECC88 (2 halves in parallel as a cathode-follower). I wonder how they do that with only one ECC88 (now we know that the ECC88 is only used in the input stage and the an opamp is used to drive the cans).

I never could compare the loudness of both drivers with each other because I did not want to become deaf. But the Futterman-circuit has the credits for sound quality. It just sounds more open and relaxing (tube-like if you will). But I have to confess that I think the guys at Musical Fidelity did a magnificent job for this kind of money and this kind of simplicity. The Futterman sounds slightly better on almost all respects than the X-Cans. The differences are not dramatic, but the Futterman sounds more tubey or lush or flowing of airy, whatever. Of course, it is all tube you hear and no IC’s in there.

Although this Futterman amp has both DC- and AC-coupled outputs, the quiescent voltage of the DC output can drift by up to 500mV, which may be too much for most headphones. The design is in principle, though, a DC design. I think that listening through the DC output is possible with higher impedance phones (which I did for many hours and nothing wrong happened), as the design is now. Furthermore, adding a simple DC-offset regulator in the feedback loop or a high voltage stabilized power supply can improve on these things.

The difference in sound quality between the MKP capacitor-coupled output and DC output is marginal. I would say that the MKP-cap adds a small effect of harshness in the highs and it robs a little of the overall dynamics and liveliness of the sound. It is better than an electrolytic capacitor though, that makes the sound ‘darker’ and more ‘shut in’, less open, more ‘boomy’ in the lows. The MKP-solution is of course an option for all capacitor-coupled amps, but do not forget that my MKP cap is only 100 Volts (there are versions of 100mF/250 and 400V but these are very very huge and pretty expensive).

Addendum

3/3/2003: Alex Cavalli submitted a version of the Stratum headphone amplifier incorporating several optimization techniques based on the work of John Broskie and circuit simulations in OrCAD PSpice. The simulation files he used can be download here. He writes:

1. The way the output of the phase splitter is connected to the SEPP stage causes the triodes to operate in grounded cathode mode (see Totem-Pole Output Stage by John Broskie). This gives the amplifier a Zo of about 300 ohms. The better way to wire the phase splitter to the SEPP stage is to cross the connections. This operates the triodes as cathode followers, lowering the Zo to about 65 ohms. This is better for low Z phones.

2. The cathode current for the phase splitter is sourced in the lower triode of the SEPP. This puts its operating point >3mA higher than the upper triode. This is not a terribly important problem, but it does limit somewhat the current swing before the amp leaves class A mode because the upper tube will cut off before the lower tube does and/or the lower tube will exceed maximum current capacity before the upper one does. So, while this is not super critical, the design is not getting the best from the output stage because it is not balanced. This fix is to source the phase splitter cathode current from the negative supply. This is done by removing the 1K resistor to ground and replacing it with a 27K resistor to the negative supply. Now both SEPP triodes have the same quiescent current.

3. The quiescent current as described in the article is probably too high. The maximum cathode current for 6DJ8/6922 is 20mA. If the lower tube is set to idle at 15mA then it can really only swing up to 20mA before exceeding this maximum rating. In the push-pull arrangement, this is a maximum of 10mA into the load (3mW peak into 32 ohms). Design maximums can be exceeded, but if they are exceeded regularly, tube life will likely suffer. Now that the idle current is balanced between both SEPP triodes, it is better to adjust both bias supplies so that approximately 10mA flows through each tube. This would be 470mV across each test resistor (with some normal variation needed to adjust the DC output to zero). This bias point permits the output stage to swing 20mA (13mW into 32 ohms) before leaving class A and without exceeding maximum ratings on the peak current. This change will also decrease the distortion some because both tubes are at the same operating point and are supplying the same amount of current thereby taking better advantage of the natural even-harmonics-cancellation of the SEPP stage.

4. The amp has a lot of gain, exceeding maximum currents with less than 0.1V at the input. We can burn some of this extra gain in a feedback loop that will lower both the distortion and the Zo. For the values shown, Zo goes to 10 ohms. For a 32 ohm load at 20mA peak, distortion goes from 1.4%% (with lots of harmonics) in the original design to 0.4% (with many less harmonics) in the modified design. The inputs are .08V peak and .25V peak respectively at 1KHz. Simulations are using PSpice 9.2. (Note that even in this example, the original design is exceeding maximum current for both triodes). Even though simulated distortion values don’t really take into account all possible distortion mechanisms, these results indicate that the changes should bring about a comparatively better distortion figure.

5. This amp can have a direct-coupled output. The same type of adjustments apply, except as I explained in my remarks that the idle current should be set for about 10mA. For example, adjust the lower bias pot until about 47mV appears on the lower triode’s plate resistor. Then adjust the upper bias pot until the DC offset is zero. This will set both tubes to about 10mA idle current. Of course, idle current in the upper tube can be measured using its plate resistor. These changes, however, won’t affect the DC drift much. But, they will help some after the tubes get broken in.

Rudy van Stratum replies:

The modified circuit is a real addition to the discussion. The text of Alex clearly shows the differences. A few important remarks however are in place:

1. Right. These Zo’s are without feedback. Remember that my design uses quit a lot of feedback and therefore has a lower Zo than 300 Ohms. One of the goals of the design was exactly copying the Futterman-patent into a headphone design, not done before to my knowledge. This includes the special way of arranging feedback via the 1k resistor.

2 and 3. Also right. I think the 470mV across the test resistor must be 47mV. And again: the difference between idle currents at bottom and top are a logical consequence of the Futterman-design.

3. The modification of Alex does away with the Futterman concept almost completely. Removing the 1k resistor is removing the feedback. Alex replaces this one resistor in effect with 2 new ones: R10 and R15 (thereby getting the output cap into the feedback loop). Alex does not mention the distortion figures and gain-factor and Zo etc with the 1k resistor added in my design. Therefore the comparison between circuits is misleading. My design has not a lot of gain, I guess that 1V of input is not a problem (comparable gain with the X-cans of MF).

In summary:

Good suggestions and probably a better overall result (to be tried by future builders).

My design has as one of the goals the replication of Futterman’s patent, therefore a slight imbalance between idle currents of top and bottom as a logical consequence. The Futterman circuits are very famous for their outstanding sound quality – see, for example, the articles and books of Rosenberg (in search for musical ecstacy, 1994).

The feedback arrangement of 1k is crucial to the Futterman design. When making comparisons please take the measurements with the feedback loop connected in my design. In effect, the solution of Alex makes it a completely new design. It has to be seen whether the differences in Zo and distortion are that great in the end and how the circuits behave in real life circumstances (of course I can not judge on that because nobody has ever heart the circuits alongside each other).

c. 2003 Rudy van Stratum.

A Low-Voltage Class-A Tube Headphone Amplifier.

by Helmut Ahammer

Warning: Like other projects using tubes for amplification, the circuits described in this article contain high voltages, and, therefore, the risk of a lethal accident is evident. Neither the author nor HeadWize is responsible for any damage or harm resulting from the construction of this project. DIYers should be familiar with and follow high-voltage safety precautions when building this amplifier.

This amplifier (which I call the “VR2”) is the follow-up of my previous HeadWize project (Tube Headphone Amplifier/Preamp with Relay-Based Switching) and has been designed with following conditions having in mind:

1) Pure Class-A triode OTL design and only one tube for amplification.
2) The plate voltages should be low.
3) The output impedance should be as low as possible and the maximum output current should be as high as possible.

This amplifier is a stand alone Headphone amplifier without the property of input selection as mentioned in my previous project but could be expanded if desired.

ad 1) Preferring the Class-A OTL design I decided again to implement a long tailed pair for the input section and a parallel connected cathode follower for the output section.
ad 2) The voltages should be as low as possible. This condition was taken into account mainly because the risk of any high voltages at the output of the amplifier, if any damage occurs, should be kept as low as possible. This could be the case if the output capacitor shortens and then the full cathode voltage would be connected direct to the output, damaging the headphone or there would be the risk of touching lethal voltages! In my opinion it’s better that this voltage is 40V and not 150V or even higher.
ad 3) Low output impedance and a relatively high current for the output sections are needed to drive headphones and in connection with the condition 1) and 2) only a few tube types are suitable.

These conditions could be fulfilled very good with the 6DJ8, ECC88 tube type family. An operating point with a plate voltage of only 80V is well between the limits and the gain µ of 33 is well enough. For the parallel connected cathode follower the E288CC is well suited, delivering higher currents and an output impedance of 25 Ohm. The voltage at the output coupling capacitor could be with this tube about 40V.

Circuit Description

Figure 1

Figure 1 shows the whole circuit of the Headphone amplifier for one channel. Even the high voltage decoupling resistors (1k5 and 330 Ohm) and capacitors (100µF/400V and 220µF/400V) are implemented for each channel separately.

The audio input is connected to a volume potentiometer (47K log, ALPS, Noble or Panasonic) and then coupled with a 220nF MKP capacitor to a long tailed pair with the E88CC triodes . The long tailed pair delivers a signal which is not phase inverted and therefore the whole amplifier is not phase inverted. Pin 7 is connected to ground, because the amplifier is designed for asymmetrical input signals. If an upgrade to symmetrical signals is desired pin 7 could be connected to the second input signal. The 1M resistor connects the grid of the first triode to ground and the 100 Ohm resistor blocks RF oscillation of the circuit. The common cathode resistor with 180 Ohm sets the operating current of 5mA for each triode and the plate resistors 7k5 set the plate voltages of about 80V. The anode of the second triode is connected to the 220nF coupling capacitor. The output section is a parallel connected E288CC cathode follower. The 470K, 33 Ohm and 1K/3W resistors set the operating point of the tubes (about 76V plate voltage and 21mA plate current). The cathodes are connected to the output coupling capacitors 470µF/400V and 1µF/400V. The output resistor 4k7/1W pulls the output to ground if no load is connected.

High Voltage DC Supply

Figure 2

The supply was build up for each channel separately, only the power transformer is used for both channels (Figure 2). To save money I decided to serial connect the secondaries of two easy gettable toroidal transformers. The primaries are connected in parallel. Two smaller transformers instead of one big transformer have the additional advantage that they can be placed very space saving into the chassis. The serial connection of 2x18V and 2x55V transformers gave under load an AC voltage of about 167V.

I used a tube rectifier for this project, because the rectifier tube implements a slow turn-on characteristic for the amplifier tubes and eliminates turn-on cracks. In my first project, I used a lot of electronics to do this job. When using tube rectification (EZ80) with only one supply voltage it is necessary to implement full wave rectification with two additional diodes (1N4007). The cathode of the rectifier tube is connected to the capacitor 47µF /450V and a 20Hy choke with at least 50mA current specification. The 47µF capacitor should be of high quality and rated at least with 400V because the periodic current loads are the highest for this capacitor. The output capacitors are a combination of electrolytic (1000µF/400V) and foil (1,5µF and 100nF MKP) types. The 220K/2W resistor should be soldered near and directly to the big electrolytic capacitor to discharge the capacitor when there is no load.

The supply has a slow turn on characteristic of about a half a minute because the rectifier tube gets conducting slowly accordingly to the warm up of the heater filament. The output voltage is about 135V under the specific load of 52mA. Especially the voltage drop of the choke and the tube rectifier is dependent on the load current. If the choke with an actual resistance of 750 Ohm is replaced by a choke with an other resistance the secondary of the power transformers has eventually to be changed accordingly. Using relatively cheap toroidal transformers, it will be not very cost intensive. Independent of the secondary voltage, the minimum VA rating should be chosen so, that there is a maximal current rating of 1A.

Heater AC Supply

Figure 3

The heater supply is simply a 6.3V 40VA transformer with two 100 Ohm/2W resistors connected to ground for hum reduction (Figure 3). Additionally for saving money, I bought a cheap 12.6V 60VA transformer for halogen lamps. Then I rewinded from the secondary so much turns that there were the desired 6.3V under load. Doing so, one have to keep in mind that the VA rating of the transformer is halved if conservatively calculated. The rewinded transformer has at least the same current rating of 60[VA]/12.6[V] = 4.76A and therefore a VA rating of at least 6.3[V]x4.76[A] = 30VA. The actual current draw of all tubes is 2.75A. Therefore there is some margin for the case of using other tubes with higher heater current demands.

Construction

Tubes:

The tube for the input section is the double triode ECC88 or 6DJ8. Better versions are the E88CC, CCa or 6922 or even with less noise the E188CC or 7308. There are many brands NOS and from current productions available. I have good experience with Philips ECC88 NOS and JAN-Philips 6922. This type of tube has µ = 33, S=12.5mA/V and Ri=2k6. Nearly equivalent is the Russian 6N1P which has slightly differing specifications.

The tube for the output section is the double triode E288CC or 8223. This tube is sometimes falsely referred as a replacement for the E88CC type of tubes. Despite that this tube has only µ = 25 the operating current must be much higher. Furthermore the inner resistance Ri = 1k25 is less. The plate voltage is about the same as for the E88CC type. For the headphone amplifier output section this tube is very well suited. Relatively low plate voltage, a current of 20mA for each triode is far below the maximum power limit and with S = 20mA/V the output impedance of the parallel connected cathode follower is 1/(2 S) = 25 Ohm. This tube is a special quality double triode with a tested life time of 10 000 hours, gold pins and with a noval socket. I used Siemens NOS and the operating point matched instantly the data sheet very closely. I can really recommend this tube.

Figure 4

If there is no possibility to get this tube, the Russian 6H30 could be used with slightly changing the circuit. Comparing the data sheets it is possible to implement this tube by changing the 33 Ohm resistor. As this triode needs about -3.4V instead of -1.4V grid voltage for an operating point of 80V and 20mA per triode the resistor should be changed to about 80 Ohms for a common cathode current of 42mA. The exact value depends on the actual tube and has to be examined by a view trials. As S=18mA/V for this tube, the output impedance calculates to 1/(2S) = 27.8 Ohm. The higher heater current demand of this tube should be mentioned too. An other possibility is the use of two E88CC tubes and therefore four triodes connected parallel together. The same current of about 40mA could be achieved and with four triodes the output impedance is calculated to 1/(4 S) = 20 Ohm. I tried it and replaced the two E288CC triodes with four E88CC triodes yielding quite the same operating point. But on the long run I think it is better to split the cathode resistors that there is again one resistor for two triodes to ensure a better equivalent current distribution through the tubes. This alternative output section is shown in Figure 4.

There are many types of the ECC88/E88CC type tubes available with probably different electrical characteristics. Therefore, if the current at the 1K /3W resistor is not about 40mA it is possible to set the operating point by changing both 68 Ohm resistors simultaneously. Increased resistor values will cause a decreased current and decreased resistor values will cause an increased current. Paralleling tubes causes an increase of the input capacitance and therefore paralleling is not very often recommended. Nevertheless I had no bad experience by paralleling tubes, there is no audible high frequency roll of or so. The MOSFET has a higher input capacitance too and therefore there are problems in high speed switching applications, but there are a lot of excellent audio circuits around.

The rectifier tube EZ80 could be replaced with the 6V4 or any other tube with the same or even higher current range. The EZ80 is rated for a maximum current of 90mA. Therefore there is a margin when using two tubes, one for each channel respectively (the current demand for one channel is 52mA) . With less lifetime there could be used one tube for both channels which I would not recommend. Using one tube for both channels the better solution would be to use the EZ81 or 6CA4 with a maximum current of 150mA. Using other tubes than the EZ80 would lead probably to an other voltage drop at the rectifier diodes and therefore the plate voltage would change or this has to be taken into account and the secondary voltage of the transformer would have to be adapted correspondingly.

Transformer:

Any serial combination of transformers could be used and as mentioned above the current rating for both transformers should be at least 1A. As transformers have lower voltages with the load connected, the right combination should be worked out by experimentation.

Capacitors:

All capacitors with values up to 1.5µF should be MKP foil types. For this types I recommend the Epcos MKP capacitors. The electrolytic high capacitance capacitors should be industrial grade types. I can recommend Aerovox BHC electrolytic capacitors. They have larger dimensions as compared to standard brand types but have less resistance and a very long life time. The high voltage ratings for the capacitors are needed because the high voltage power supply is not regulated and with less load current the voltage raises. For instance, if one output tube is disconnected the voltage increases to over 200V!

Resistors:

All resistors should be of metal film type with a power handling capacity of about 0.5W or higher which is then specified in the circuits.

Choke:

At least 20Hy are recommended and the current rating should be at least or in the best case 50mA. If a choke is used with an actual current which is less than the nominal current the inductivity is decreased. The resistance of the choke determines the voltage drop when used with a specific current. If the choke has an other resistance as about 750Ohm, the high DC voltage would be changed or to circumvent this, the secondary voltage of the power transformers should be changed accordingly. I bought the chokes at the German transformer and tube gear seller Welter Electronic located in Ulmen/Eifel (Type: Dr.7 20Hy, 50mA, 720 Ohm). Sowter (www.sowter.co.uk) sells the type CB25 with 20Hy, 50mA and 451Ohm. Hammond (www.hammondmfg.com) sells the type 193C with 20Hy, 100mA and 181 Ohm.

Realisation:

The pictures shows the final realisation with a similar design as my previous amplifier. As I stated in my previous article I like to see the tubes but don’t like to see the transformers and capacitors. Therefore all the transformers and big capacitors are packed into the back part of the chassis. The chassis measures 435mm x 319mm x 122mm and is made from 1.5mm aluminium plates with enlarged hardness. The single aluminium plates are mounted together with L-shaped aluminium profiles and screws. The top of the actual version of the amplifier is built from three pieces of birch plywood which where glued together and sprayed with the same colour as the front panel.

One picture shows the same amplifier with an alternative top made of transparent lacquered beech plywood. The front panel is again made from white clay and has a thickness of 15mm. I drilled the holes after the drying period of about 6 weeks and fired it at about 1000°C in a kiln. After lacquering I engraved the letters with a high rotation mini drilling machine. Because of this ceramic front panel the amplifier is really heavy.

Results

Using good components this headphone amplifier is a high grade amplifier incorporating a really Class-A operation with all it’s sonic benefits. The use of only one amplifying device strengthen this approach. The sonic quality of this amplifiers lies in the excellent reproduction of the music especially with moderate volume levels. All the details are there without being overwhelmed by any special sonic property. Therefore this amplifier is very neutral but without any insistent behaviour. Long term listening with the Sennheiser HD580 or the Beyerdynamic DT770 Pro (250-Ohm) headphones is really a pleasure.

The overall gain = 12dB and with modern CD-players, which have outputs of 1V up to 2V the gain is far enough to get really loud volume levels. With my Philips CD-Player I turn most of the time the volume at the position 1/4 from maximum and the position 1/2 is really loud. I can’t hear with the maximum setting because this is far too loud and ear splitting for me. If really more gain is necessary there is the possibility to connect a capacitor (10µF 200V) to the 120V connection and the pin 1 of the E88CC triode. This capacitor shortens the 7k5 resistor for AC voltages and increases the gain of the first triode circuit which is principally a cathode follower. This capacitor should lead to a gain of 18dB. More gain is even achievable if the E88CC is not wired as a long tailed pair. Instead of the long tailed pair it is possible to build up two common anode amplifiers using separate cathode and anode resistors and an additional coupling capacitor. In this case the phase isn’t inverted too.

Compared to my first project I can say, that the bass control is increased (mainly because of the lower output impedance of 25 Ohm). Especially the bass with the DT770pro is more defined and accurate, but this behaviour is less pronounced with the HD580. Overall the tonal behaviour is very neutral and balanced with a slight bit of warmth and it is not the type of amplifier which is clinically detailed. Compared to the headphone output of the Philips CD-player, the tonal presentation is full of life with a very clear representation.

Measurements:

Finally here are some values from measurements I have done so far:
Frequency response: 10Hz-100kHz
Overall Gain: 4
Output Resistance: 25 Ohm (24.8 Ohm and 26.7 Ohm for both channels respectively)
Maximum Output Current: 300 Ohm or 30 Ohm load: 23.3mA rms, 33mA peak
Maximum Output Power:
300 Ohm load: 163mW rms, 327mW peak
30 Ohm load: 16.3mW rms, 32.7mW peak

c. 2003 Helmut Ahammer.

A Tube Headphone Amplifier/Preamp with Relay-Based Input and Power Switching.

by Helmut Ahammer

[Warning: Like other projects using tubes for amplification, the circuits described in this article contain high voltages, and, therefore, the risk of a lethal accident is evident. Neither the author nor HeadWize is responsible for any damage or harm resulting from the construction of this project. DIYers should be familiar with and follow high-voltage safety precautions when building this amplifier.]

Tube amplifiers designed for headphones have the principal property that they can be used as preamplifiers too. In most cases, the output impedance of a tube headphone amplifier is (or should be) less than the output impedance of a tube preamp. Given this advantage, the use of a tube headphone amplifier for preamplification is not so critical concerning the input impedance of the power amp or cable impedance and cable capacitance. This project expands on the basic amplifier, with a slow turn-on for tube heater and plate voltages and input section using high quality relays.

The main intention of the project was the minimal use of elements in the audio path and the creation of a fully-featured amplifier with high audio impact. Therefore, additional discrete semiconductors and digital (logical) integrated circuits were brought in, serving only as helping devices. Only one tube in the audio path amplifies the audio signal. Here I have to say that I prefer the pure Class A topology, using only a single amplification stage with a parallel connected-output triode without overall feedback. This topology is not new and is for instance also mentioned in the Top-Level OTL Tube Headphone Amplifier and No-Compromise Tube Headphone Amplifier by Andrea Ciuffoli.

This amplifier features relay-based input and power switching. The stepped and slow turn-on for the power supplies result in less stress for the tubes and other components and reduces turn-on thumps that could damage headphones. Most tubes fail at turn-on. Without the slow turn-on, if the tube heater filament is cold, the resistance is lower and the in-rush current of a cold filament could be very high and cause the filament to break. And if the filament is not hot, it is better that the plate voltage is not applied. Applying the plate voltage with cold filaments can reduce the lifetime of the tubes.

I prefer relay input switching over input switching with a rotary switch for two main reasons. First, it is very hard to get a really good rotary switch, and when available, it would be very expensive. In contrast, there are many relays on the market, the contact material is normally very good, and they are relatively cheap. Second, it is possible to locate each input relay very near to the corresponding input jacks. Then there is the big advantage, that the wiring from the input to the switch is very short, and only one stereo cable must be routed through the amplifier. If the tape loop is used, then there is also a very short wiring for the audio path.

The circuits in this article are presented more or less as independent from each other, providing the possibility of an easy change if desired. The headphone amplifier could be built without the preamp sections (input relays and so on) or any part of the power supply could be changed (for example, valve rectified) or a preferred amplifier topology not presented here could be extended with the preamp sections.

CIRCUIT DESCRIPTION

Headphone Amplifier/Preamp

Figure 1

Figure 1 shows the circuit of the tube amplifier designed without global feedback. It is a pure Class A OTL design with a triode-connected pentode output section. The input section was built with the double triode E83CC as a differential amplifier. The big advantage of this circuit is that the output signal can be taken from the first or from the second triode. The output at the plate of the second triode isn’t phase inverting providing that the whole amplifier isn’t phase inverting. The E83CC is very often mentioned as a tube with excellent audio properties. There are selected types and different brands on the market. The E83CC is the high-grade type of the ECC83. It has a very high gain (µ = 100) and is therefore well suited for building up an amplifier with only one amplifying device.

To circumvent having a high voltage negative power supply for the current source (simply R4) of the differential amplifier and for improving the linearity, it is necessary to lift the cathode voltage and the grid voltage of V1 above zero volts. This is done by R7 and R8 acting as a voltage divider. This voltage divider could be built two-fold, each for one of the triodes, but since tubes have a negligible input current (gate current), the same divider gives the gate voltage for the first triode too through R6. With a gate-cathode voltage of about -1.2V and R4 = 4.7K Ohm, the plate current for each triode is about 1mA. This current increases the lifespan of the tube and gives very good results. The resistors R2 and R3 are the plate resistors and provide a plate voltage of about 170V (with a 280V power supply).

The higher the power supply voltage the higher the obtainable gain. With a power supply of 280V, the gain of the input section was measured to be about 22dB. R1 and R5 are gate resistors commonly used to reduce RF oscillations. Because the differential pair only uses one input, the other input must be grounded. C2 grounds the second input at audio frequencies. The gate is at a lifted potential for DC voltages, but is at ground for AC voltages. Like the input decoupling capacitor C1 and the interstage coupling capacitor C3, the gate capacitor C2 should be high performance too. Industrial standard MKP types (e.g. Epcos or Wima) are recommended.

The output stage is a cathode follower (gain < 1) with a triode-connected pentode EL84 instead of paralled triodes. If I had used triodes with high enough current (30mA or more), the plate voltage would have been, say, about 100V-120V. Then the voltage at the cathode resistor would have been in the range of 160V-180V. As the maximum cathode heater voltage must be less than 100V or in practical terms less the 80V, the ground plane of the heater supply would have to be lifted by 80V-100V. But lifting the ground plane with all the implemented control circuits (relay control, CMOS circuit, etc.) is not good. Furthermore, I dislike high voltages at the output capacitor. A fault of this capacitor could pose a lethal risk. Therefore I decided that the cathode resistor voltage should be not more than 80V. But what to do with the remaining 200V! There are no noval socket triodes (relatively cheap triodes) that can withstand this high plate voltage.

The solution I found was the triode-connected EL84 with a 210V plate voltage and about 35mA current – permitting a cathode resistor with lower wattage and lower cathode voltage! The drawback is the higher output impedance compared to paralleled triodes. The triode mode of the EL84 has lower output impedance and distortion compared to the pentode mode. R11 connects the second grid to the anode for the triode mode. The cathode current of 35.5mA is set with the cathode resistor R14 and the biasing resistors R12 and R13. The paralleled biasing resistors could be changed to one resistor with a value of about 180 Ohm and with a higher power handling capacity.

The output impedance of the cathode follower is approximately calculated with Rout = 1/S (S is the transconductance of the triode connected pentode). From the data sheet plate current/ gate voltage graphs in triode mode a value of S= 16mA/V, slightly higher than 12mA/V for the pentode mode is arrived and therefore Rout calculate to about 60 Ohms. This value works fine with headphones of 300 ohms impedance concerning the damping factor. By using headphones with about 30 Ohms, not only there is no damping, but furthermore the gain of the stage would be reduced. If gain isn’t that of importance low impedance headphones could give good results too, despite of the lack of damping.

Possible changes:

The resistor R4 with an actual value of 4.7K Ohms is not a real current source as the impedance of a current source should be much higher (about 1M Ohm). Implementing a better current source, it is possible to replace R4 by a choke or a FET current source. Using a BF245 with a resistor connected between Source and Gate could be replaced with no change for the rest of the circuit. A potentiometer could make the current adjustable. Bypassing R2 by a capacitor increases the gain of the input stage and would lower noise of the circuit. With the experience of the used power supply, which is very stable, noise was no problem and the gain without this capacity is well enough for driving 300 Ohm headphones.

With low impedance headphones, say 32 ohms, the output tube could be used with a current of up to 49mA instead of 35.5mA but then the voltage of the tube, the cathode voltage or the power supply voltage have to be changed. The impedance at the output is connected in AC terms parallel to the cathode resistor and the gain is reduced. Therefore, the maximum voltage swing is limited, and the current through the load is less. Adding an output transformer with a high primary impedance and a low secondary impedance could solve, but it would be better to integrate a transformer directly to the anode or cathode of the tube. Personally I prefer the transformerless approach, because these transformers need to be in the most cases very special ones and therefore are very expensive.

I thought of a using a real Class A MOSFET Follower instead of a tube output stage. Combining the gain (and the sound) of the E83CC tube with a MOSFET (low impedance) output is very interesting but actually I haven’t built such a circuit.

Heater DC Supply with Slow Turn-On

Figure 2

To minimize the possibility of hum problems (arising from AC heating of the tubes), a regulated power supply is used. Furthermore a slow turn-on was implemented (which could be found in similar form in many textbooks about tube circuitry) to limit the big in-rush current (figure 2). This slow turn-on increases the lifespan of the tubes. C1, C2, C3 and C4 reduce spikes from the rectifier diodes. Instead of the four capacitors (C5, C6, C7 and C8), one big capacitor could be used but several capacitors decrease the ohmic losses. The adjustable voltage stabilizer LM317 is used with the diodes D5 and D6, which are safety diodes in cases where negative voltages occur (mainly from turning off the amplifier).

The combination of R1, R2 and P1 set the output voltage. The combination of R4, C9 and Q1 gives the temporal behavior of the circuit at turn on. At the first moment C9 isn’t charged. Charging the capacitor through R4 gives a relative high voltage at this resistor and therefore Q1 is fully switched on. Q1 gives a very low parallel resistance and a very low overall resistance for the LM317 and therefore a very low supply output voltage. By charging C9 the overall resistance for the LM317 and the output supply voltage are raised. The output voltage before diode D7 is about 13.4V (the actual value is not critical) and is for the relay and control sections. The heaters of tube V1 (E83CC) of the left and right channel are parallel-connected. The heaters of tube V2 (EL84) for the left and right channel are connected in series, because they work only with 6.3V.

High Voltage DC Supply with Stepped and Slow Turn-On

Figure 3

The high voltage supply is turned on after a 40-second delay through a current-limiting resistor for slowing down the voltage ramp-up. For the first 40 seconds after turning on the amplifier, only the heater and the relay and control sections are powered. There is no voltage at all at the plate of the tubes (e.g., the output voltage of this supply is 0). After this 40 seconds, the control circuit turns on the relay Rly1 and the power supply for the plates begins to work (figure 3). As the MOSFET (Q2) configuration has also a slow turn-on function, the plate voltage rises slowly and, therefore, the tubes are not stressed. After 60 seconds, Rly2 shorts R1 and the voltage increase is accelerated. R1 and the contact of Rly2 could be placed at the primary of the transformer to limit the in-rush current of the transformer too.

The transformer is an easy-to-get 1:1 or 1:2 (depending on the primary voltage) transformer and should have a power rating of at least 100VA. R2 and R3 effectively increase the impedance of the rectifier diodes. This mimics in a minor manner a tube rectifier because tubes have a higher intrinsic impedance when compared to rectifier diodes. C1 and C2 minimize voltage spikes originating from the rectifier. C3 is actually built from six 150µF capacitors providing fewer losses. R4 and R10 are directly and closely connected to the capacitors to minimize the danger of charged capacitors if there is a fault (e.g., loosened connections to other circuit parts). Q1, D3, D4, R5 and DZ1 set the reference voltage of 285V, which is further smoothed by C4, R7 and C5.

Connecting the headphones (300 Ohms), there is absolute no audible hum or noise, except the volume pot is at the position 9 of 10. At this position it is able to hear the amplified input fluctuations but nevertheless no hum from the power supply. R9 is a safety resistor, if the connection to the output capacitors C6 is broken.

Relay-Based Input-Output Switching Section

Figure 4

The input-output switching section with input resistors for decoupling, a tape loop function (Rly7), the volume pot P1 (high quality recommended), the amplifier itself, an output relay Rly8, decoupled line outputs and the headphone output are shown in figure 4.

It is not necessary to buffer the inputs. There are different resistor values for the decoupling resistors mainly because today’s CD Players have 1-2Vrms outputs and many other inputs (like tuners, etc.) have outputs with less voltage. Therefore, I decided to halve the output voltage of the CD player, as then there is only a small change of volume when the inputs are switched between the CD player and an other units. The resistor values for the CD-input are relatively low because the output impedance of a CD player is normally relatively low too. The resistor values for the rest of the inputs are relatively high (the resistors to ground) because many tube amp owners have other tube gear to be amplified. For this case, the resistors should be not too small as the output impedances are normally higher. Principally, high resistances have the drawback of producing more noise, and the actual values of the decoupling resistors do not have to be exact.

The tape loop could be used to implement a signal processing circuit like EQ or crossfeed when using the amplifier for headphones. The input relays are switched by darlington transistors (BC517). The electrolytic capacitors (C7 – C10) minimize switching pops or crackles.

The tape loop relay (Rly7) and line output relay (Rly8) could be turned on with a normal switch, or with a momentary switch in combination with a NAiS VS5-24V electronic switching circuit and a bistable relay (Rly9 and Rly10) as shown in figure 4. The VS5-24V integrated circuit and a cheap polarised 2-coil relay (from any company) has the function of an expensive and hard to find stepper relay (a relay with two stable switched positions set by pulses).

The same circuit is used for the line output relay (Rly8), but in this case the relay can be switched only after the whole turn-on delay of 70 seconds. When the Tape/Line momentary switch is pressed, the second switch contact of Rly9/Rly10 (between the coil of Rly7/Rly8 and ground) is switched permanently. The other switch contact of Rly9/Rly10 between the two coils of Rly9/Rly10 is only necessary for the proper function of the VS5-24V. LEDs on the front panel indicate the status of Rly7/Rly8.

Turn-On Delay Control

Figure 5

Figure 5 shows the circuit for the turn-on delays which control the course of the turn-on procedure. The CMOS circuit 4060 is a binary counter and Q14 is the highest bit. C3 and R3 give the frequency of the oscillator and therefore the delay duration. At the start, C1 is not charged and pin 12 (RESET) of the 4060 is HIGH and resets the counter. Charging C1 sets pin 12 LOW because of R1, preventing a fault reset. The first time Q14 goes HIGH (after 40 seconds), Q1 is turned on. If, in addition, Q13 goes high (after 60 seconds), the NAND gates IC2C and IC2B turn on Q2. And finally if, in addition to Q14 and Q13, the pin Q12 goes HIGH (after 70 seconds), the gates IC2D and IC2A turn on Q3. D1 stops the oscillator to preserve this state. Changing the oscillator frequency would change the total duration of the whole delay, but would preserve the proportional timing. The formula for setting the frequency is: f = 1/(2.3 * R3 * C3). C1/R1 are reset at power-on.

This circuit starts every time in the same manner, and all the steps are cycled through with the depicted timing, regardless of the usage of the main switch (for example, if the main power switch is turned off and on improperly).

Flashing Led Signaling Slow Turn-On Procedure During Power-On

Figure 6

The circuit in figure 6 provides a rough visual display of status of the different steps of turn-on (and therefore representing the state of the amplifier during power on). Q1/Q2, D4/D5, C1/C2 and R2/R5/R6/R9 are the common components of an astable oscillator circuit. When the amplifier is powered on, the LED (D3) flashes slowly with the time constants given by C1, C2, and mainly R5 and R6. There is no second LED in connection with R2. This asymmetry ensures that the circuit starts oscillation under all circumstances.

After 40 seconds, the turn-on delay control for Rly1 (figure 3) also turns the transistor Q4 on. Then R6 is paralleled with a low resistance and gives a shorter time constant with C2 and, therefore, a faster flashing rate for the LED. Similarly, after 60 seconds, the Rly2 (figure 3) and the transistor Q3 are turned on and the LED flashes again faster. The diodes D1 and D2 prevent the toggle circuit from incorrectly turning on the relays. Playing with the values of R4, R7, C1 and C2, it is possible to change the flashing intervals of the LED. The depicted values are given for operation where the no-light durations are smaller than the light durations.

D1 and D2 prevent the toggle circuit from incorrectly turning on the relays. A “low” signal, produced periodically by Q1/Q2 could eventually be conducted through C1/C2, Q3/Q4 and R3/R8 to the coils of Rly1 and Rly2 during the delay time. Probably the resulting currents to the coils would be far too small to really turn on the relays, but I wanted to prevent it in any case.

And finally after 70 seconds, the output relay is turned on. Q5 and R1 stop the toggle function of the circuit and the LED operates in continuous mode. Overall this gives a very convenient optical control of the start-up procedure with one LED, which is located at the very left side of the faceplate directly above the main power switch.

CONSTRUCTION

The 6BQ5 and The Russian 6P14P (6P14P) in figure 1 can be substituted for the EL84 with very good results. The decoupling output capacitor (C4) should have at least 220µF with an appropriate voltage rating. A high voltage rating with a big amount of margin for this cap lowers the possibility of a failure (70V DC at the output!) and is therefore strongly recommended. Raising the value of the capacity gives better low frequency results. Bridging with an 1µF MKP type is recommended. All resistors are of 0.4W-0.6W type if not otherwise specified in the schematic.

In figure 2, if the BC557 transistor is not available, try 500mW types such as the BC556, BC327 and even the 300mW types like the BC177 should work as well. I’m not aware that there is a direct substitute for the ZTX758 in figure 3, because few PNP types are rated at 400V. If the ZTX758 is absolutely not available, then I would recommend 300V types such as ZTX757 or MPSA92. Substitutes for the IRF840 MOSFET include the IRF740, IRF830 or BUZ40B or any other N-channel MOSFET with a rating higher or equal than 400V, 4A should work. Types with less R_dson should be preferred. The fuse labelled 75 degrees (C) is a temperature fuse mounted inside the chassis which cuts the high power supply, if the temperature of 75° is reached. There are no direct substitutes for the BC517 darlington transistors in figure 4, 5 and 6, but the BCX38C could be used. For the BC107 transistor in figure 6, any small NPN transistor like the BC337, BC546 or 2SC1815 would also work.

The MOSFET IRF840 (figure 3) needs a heatsink with a temperature resistance of less than or equal to 6K/W, and the voltage regulator LM317 (figure 2) needs a heatsink of less than or equal to 4K/W. The “hot” parts of the amplifier (tubes, heatsinks, ev. transformers) should be equally distributed in between the chassis. Mounting the heatsinks in a way that they are an outer part of the chassis should be preferred.

At this time, there is really no information on the net about the NAiS VS5-24V module. I have a single data sheet from the local distributor. RS components and some local distributors here in Europe (e.g., Schuricht Elektronik) sell it. Furthermore I bought a small quantity (2 pieces) direct from the local NAiS distributor. The price is about Euro 9.30. There is no substitute, but I think it would be not so hard to develop a discrete version by any advanced DIYer. “Aromat” is the USA brand for NAiS, but NAiS does not know if Aromat sells the module. If not, it could be purchased from NAiS in Europe. I recommend contacting Aromat about the module and the availability.

If it is impossible to get the VS5-24V module, omit the Rly9/VS5-24V combo and replace the switch SW9b with a push button single pole switch. Electrically this switch could be a cheap one, because neither audio signals nor high currents nor high voltages would be switched. The same for Rly10.

All of the relays should have 12V coils with a resistance greater than 700 ohms. Rly1 and Rly2 have to be good power relays, such as the Siemens V23092 (any reliable monostable relay rated for 230VAC, 4A-6A works well in this application but cheap products should be avoided). They need only one switching contact (SPST). The switch contacts of Rly1 and Rly2 should NOT be moved after the rectifier to switch any high voltage DC – they should switch only high voltage AC. Normal “230V” relays are not constructed to switch 230VDC. I once had a bad experience with a relay specified for 320 VDC to switch 280VDC. The contacts melted together!

The use of high quality 12V relays for Rly3 – Rly8 is highly recommended (e.g., P2 Siemens or DS Nais DPDT types). For switching audio signals, it is important that the relay has a low and constant switch contact resistance at low voltages and currents. The resistance of a relay switch contact is specified with a minimum and maximum rating (the minimum rating should be as low as possible). For example, the Siemens P2 relay has a minimum voltage rating of 100µV (modern power relays (e.g., Siemens V23092) have a minimum voltage rating of at least 5V). The bipolar relays (Rly9/Rly10) have DPST switches. They do not have to be of the same quality, because they aren’t in the audio path and don’t switch high voltages or currents.

The part number of the Siemens P2 relay is V23079. DigiKey in the U.S. sells a Potter and Brumfield V23079 relay that is the same as the Siemens P2 relay. All Siemens relays have this type of number code. Despite that there are slightly different dimension values, I think it is really the same relay, especially since the coil current is exactly the same. The bistable relays Rly9 and Rly10 can be cheap products, because they don’t switch high power and don’t switch audio signals. They must be polarized, 2-coil types and must have at least DPST contacts.

Using a high quality momentary switch for the Tape and Line controls results in a very professional feel. I used the Apem type 18535CD, which costs about Euro 5.84.

As can be seen in the picture at the top of this article, this project could be built up in a relatively small case. The front plate was made of a 15mm ceramic plate. Ceramic is very heavy but absorbs vibrations very well and is therefore well suited for audio gear. The surface is manually fire lacquered giving a slight structure to the appearance. This plate is not 100% flat and uniform but therefore reflects on the other side the hand made aspect and individuality and completes for my opinion the tube approach very well.

To make the faceplate, I used white clay which deforms not so much when fired, and pressed it into a wooden form and dried it for about 4 weeks before firing. The wood form is a piece (about 40mm) of glue-pressed wood with material cut out with a router. The faceplate was fired in a kiln slightly above 1000°C for end stability (only drying the clay does not work). After firing the faceplate, I lacquered it with common spray lacquer (I didn’t use any glaze). The lettering is engraved with something like a small drilling machine and with high rotation. The white colour is the white clay under the lacquer. I tried drilling holes for the switches and pot in the faceplate both before and after firing and had different success, but actually I don’t know which method is better.

The chassis measures 435mm x 319mm x 142mm and consists of single aluminum plates mounted together simply with L-shaped aluminium profiles and screws. The thickness of the aluminium should be at least 2mm, and there should be enough holes in the chassis for cooling the components. Apart from that, many DIYers like to see the transformers and the big capacitors. I don’t, and therefore I constructed this chassis, where only the tubes are visible. The top plate extends over the tubes for protection. To mount the ceramic faceplate, I drilled little holes in the backside of the faceplate and glued little threaded bolts (threaded rods) into it. Then I mounted it with nuts to the aluminium chassis.

RESULTS

For testing, I would recommend measuring the voltages across the cathode resistors as they determine the current through the tubes (see the DC operating points shown figure 1). Also measure the heater voltage, the high power supply voltage and the voltages at the anodes. If all these voltages are OK, the amplifier should work properly. The start up procedure could be tested easily by hearing if the relays click at the right order in time. The proper function of relay Rly1 and Rly2 could be tested further by measuring the voltage right after the switch contact of Rly2 (before the rectifier). The slow turn-on of both the high voltage supply and the heater supply could be tested by measuring the voltages during the start up.

The voltage drop at the MOSFET Q2 (high voltage supply) should be in the range of 20-25V. When all the tube heaters are connected and warmed up, the output of the low voltage (heater) power supply is set to 12.6V with P1. When the power supply is turned on for the very first time, it is recommended to set the trim pot to the mid position and then adjust for 12.6V at the output. After connection of all the loads and after warm up, the voltage must be readjusted.

The heaters of V2 are connected in series across the 12.6V supply. If there are big differences in the heater resistances, it could be that one heater would run with a higher voltage. If each individual heater voltage does not exceed the specification (mostly ±10%), minor differences don´t matter. I have never had this problem, because normally I use pairs of tubes with the same “history,” and even during experimentation with different tubes (age, brand, etc) I had no problems. It could be, that even a tube from the same batch has a failure, and then the second tube would be influenced concerning tube performance or lifespan. Therefore, I recommend checking the heater voltages, when the tubes are turned on the first time.

Here are the output current measurements of the amp driving a 30-Ohm load and a 300-Ohm load:

30 Ohm resistance:
Maximum current 15.5 mA rms without clipping
Maximum current 32 mA rms with clipping

300 Ohm resistance:
Maximum current 10.5 mA rms without clipping
Maximum current 20 mA rms with clipping

I have compared this headphone amp to an old Technics receiver and to a hand-made solid state pre-amp built with a NE5534 opamp and a push-pull BD139/BD140 transistor output stage for driving headphones. The Technics receiver is the worst because of noise at the output and inferior audio quality. Mainly, there was no clear musical representation and therefore a very bad localisation of single instruments. My hand-made amps perform, in my opinion, very well. It’s hard to say why, but I like my tube amp more because of the overall tonal representation, especially during longer listening sessions. This is the reason why I presented my tube headphone amp and not my semiconductor amp. I’m using only my HD580 with this amp.

c. 2002 Helmut Ahammer.

The 6N1P OTL Headphone Amplifier.

by Bruce Bender

The HeadWize DIY forum is full of information about opamps for use in portable headphone amplifiers. The Pocket Amplifier by Chu Moy (a.k.a. the “CMoy amp”), especially with Carl Hansen’s circuit board, is a great project. I built a few of those, but for my next project, I wanted to go a bit further and do something a little more discrete. Portability was not an issue for me, since most of my headphone listening is done at home in the living room where my stereo lives.

I have had the pleasure of using tube amplifiers and tube preamps of various types over the years, and I do believe that, all else being equal, tubes sound better than transistors. So I wanted to build a tube headphone amp. I also wanted a simple amp, especially the power supply. In a lot of tube projects the power supply can be more complicated than the audio circuitry. After reviewing the various tube headphone amp designs on HeadWize, as well as some circuits from other sources, I chose the Morgan Jones amp. I thought it would be a lot of fun to build an amp that uses only one kind of tube, to really get the flavor of that tube, and I also was interested in the output-transformer-less (OTL) circuit, with no global feedback.

I was intrigued by the Svetlana 6N1P tube. I much prefer to use tubes that are still being manufactured, having had very mixed results with new old stock (NOS) tubes. Svetlana was promoting a large lineup of tubes in the USA at the time, the 6N1P being one of them. The 6N1P has good performance curves, and supposedly is compatible/interchangeable with the 6922 family, although my experience seems to show otherwise. This amp is really optimized for the 6N1P tubes, so if you want to use 6922s you should use the circuit in the Morgan Jones article.

I also realized that this would be a good first tube project for HeadWizers that want to venture out from the world of opamps into the world of tubes. So I decided to document the project with narrative and a few pictures, to make it as straightforward as possible for anyone else to try it.

Like the original Morgan Jones amp, the original 6N1P OTL had trouble driving low impedance headphones. This version of the 6N1P OTL amp features a re-balanced output stage that gets all of my headphones plenty loud. Alex Cavalli ran PSpice simulations to get the new parts values. The rebalanced amp also needed a higher voltage power supply. For more information on the Morgan Jones amplifier or theory behind the optimization, see The Morgan Jones Mini Tube Amplifier. For more information on the basics of tube electronics, visit the Svetlana or Triode Electronics websites.

The Amplifier

Figure 1

The amp is built according to the Morgan Jones design. In the spirit of DIY, I made a few modifications. I did not change the circuit topology at all, but used Svetlana 6N1P tubes, as noted. I also used a conventional power supply instead of the wall wart/filament transformer combination, because the 6N1P has a higher filament draw (2 amps) than 6922s, and I wanted plenty of high-voltage supply, too.

If you can read an op-amp circuit diagram, you should be able to read this tube circuit with no trouble. The 6N1P family are “dual triodes,” each tube containing two separate amplifiers in the same device. Each triode has three circuit connections: a source of electrons (“cathode”), an output that receives the electron flow (“anode” or “plate”), and a “grid” which controls the flow. A separate wire filament (the orange glow in tubes) heats the cathode to make it give off electrons.

Figure 2

Figure 2 shows the tube pin layout. The layout is shown from underneath, from the bottom of the tube. This makes sense, since that is the view you have of the tube socket when wiring it. Pins 1, 2, and 3 are the anode (A), grid (G), cathode (K) for the first triode (T1) inside the tube. Pins 4 and 5 are for the filament heater (H). Pins 6, 7, and 8 are the anode, grid and cathode for the second triode (T2). Pin 9 is for shield (S) between the two triodes in the tube, and in this amp should be connected to ground. In the schematic, all the cathodes are shown on the bottom (the little hat shape signifies the heater filament), grids in the middle, anodes on top.

The Power Supply

The power supply shown for the original Morgan Jones amp is unusual and clever, but the 6N1Ps need a higher voltage for the plates and more current for the filament heaters than the original supply can produce. [Editor: The 6N1P needs a higher plate voltage than the 6922/6JD8 to reach its optimal operating range.] The heater current is 600mA for the 6N1P versus 365mA for the 6922, a significant difference. The clever wall-wart scheme, which is barely adequate for the 6922s, clearly can’t provide enough current for 6N1Ps. So I used a more traditional power supply. Tube amps traditionally have a power transformer that has a separate winding for the heater filaments in addition to the main winding. As it happens, a transformer is available that provides usable voltages for this project, the Hammond 269AX.

Figure 3

Figure 3 shows the power supply. The high voltage supply is 350V. Since the cathode-heater voltage of the top half of V2 then exceeds the 6N1P’s spec of 100V, I floated the heater supply ground with a 0.22uF, 400V capacitor, so the whole circuit connects to ground only via this grounding capacitor. I don’t know if 0.22uF is an optimum value for this use – I just had one on hand.

Do not use the center tap (red/yellow) on the high voltage secondary winding of the 269AX. Also, be careful if you run this power supply for very long without a load on it, since voltages will quickly accumulate into the 400 volt range. The amp will draw about 27 milliamps of current. If you want a dummy load to use when checking the power supply, you will need to cobble together a 13k ohm load capable of dispersing 10 watts.

The power supply output network has one 1K-ohm/2W and two 1K-ohm/2W film resistors in parallel. You can trim the output voltage by adjusting the value of the second 1K-ohm/2W pair slightly up or down from 500 ohms.

The first version of the power supply used A.C. straight out of the transformer to heat the filaments. The amp sounded fine, but had a very loud hum that didn’t go away until the filament circuit was rectified as shown in the schematic.

Construction

With opamps, voltages are rarely above ±15 volts. With tubes, voltages are much higher, up to 1,000 volts. This project has voltages as high as 400 volts. This is serious stuff and needs to be treated with the respect it deserves. There are three basic rules when working with high voltages.

• Keep one hand behind your back. A very dangerous path for high voltage is from hand to hand, with your heart in between. It doesn’t take all that much to stop your heart at these voltages if the timing is wrong.
• Use proper probes. About the only thing you should be doing to a live circuit is checking voltages or signals. Get clamping tips for your probes. Clamp the ground clip to ground. With the hand that is not behind your back, use the proper insulating probe tip on your voltmeter or scope. Be sure your voltmeter or scope is rated for at least 500 volts for this project.

  
  Figure 4

• Disconnect the power and DISCHARGE THE CAPACITORS before doing anything to the wiring. The capacitors in this project can give you a high-voltage jolt at much more than 27mA, and can hold that charge for a long time (e.g., an hour or more). Do NOT discharge the capacitors with the crude method of shorting the positive side of the caps to ground with an insulated screwdriver. I blew out a rectifier when I tried that. A better method is to bend a 10K-ohm/2W resistor into a “U” shape, tape the body of the resistor to the end of a popsicle stick, and use that to discharge the capacitors (figure 4). Measure the capacitor voltage with a meter to be sure it has drained all the way down: they don’t discharge all at once, it takes 15 or 20 seconds.And don’t forget to discharge the audio output capacitors, too!

In the fine tradition of many tube amps, this project uses point-to-point wiring (meaning there is no printed circuit board or perf board involved). Most connections are made with solid 22 ga. copper wire, which stays in place after soldering, and does not flop around as stranded wire does. I also use ordinary solder from Radio Shack. To get good solder joints, you need to firmly connect the wires to be joined in a good mechanical joint, and then solder them. Keep everything dead still until the solder cools, or you may get a “cold” solder joint, which won’t work well. If in doubt, just re-solder the connection.

Download parts list for 6N1P OTL Amplifier

The building process for point-to-point wiring is different than when you are using a pc board or perf board. Instead of stuffing the board with components, and then putting it in a case, you start with the case and attach things. There is no “right” way to do this, but what follows is what works for me.

• Mount the physical parts on the chassis first.
• I used a steel chassis. Lots of tube equipment have been built this way, with the transformer and tubes on top of a steel chassis and the wiring inside, and that’s what I prefer. Other folks seem to like aluminum, which they find easier to machine, but I don’t see that much difference myself. At any rate, do not use plastic: you can’t ground to it and it can’t handle the heat from the tubes.
• This is probably the hardest part of the project. Making big holes in 20 gauge steel is tough. You need high speed drills of the proper sizes, and preferably a drill press. Center-punch each hole center before drilling to keep the drill bit from wandering until it takes hold. For holes bigger than 13mm, I used a nibbler ($10 from Antique Electronic Supply or Radio Shack) and patience.

A standard configuration for hard-wired audio tube amps is to put the transformer(s) and tubes on top of the chassis, and the wiring underneath/inside the chassis. I put the AC receptacle and input jacks on the rear panel. The transformer goes on top, in the back right corner (don’t forget the holes for the wires) and the tube sockets go as close to the front as practical. Locate the tube sockets far enough back from the front panel so you can wire them with plenty of clearance for the volume control and power switch. The power switch, headphone jack, and volume control go on the front. (Personally, I like to have volume controls on the front panel, not on top.) Position the input jacks (in back) and volume control (in front) to the left side, as far from the power supply as possible. The power switch goes on the right.

Figure 5

To wire the tube sockets, I wanted a physical wiring guide. I printed a piece of paper with three copies of the tube pin layout and then penciled in the various connections. I tried several layouts until I found one I liked. Shown above (figure 5) is my wiring guide for an earlier version of the amplifier.

Once you have made all the holes, file down any rough edges. Test fit all the pieces. Drill pilot holes for the mounting screws. Test fit each component. Then, when everything fits, stop and paint the chassis and cover before mounting the hardware.

If you use the Hammond chassis in the parts list, it is already painted grey. I marked it up doing the layout for the parts, so I roughed it up with sandpaper and painted it. I used “American Accents Hunt Club Green Satin” by Rustoleum and do not especially recommend it, since it looks a bit like plain old olive drab. After painting, you really should let the paint dry for two days. Otherwise, it scrapes off easily. Once the amp is up and running, it gets warm enough to “bake” the paint somewhat, and you can smell the paint baking for the first week or two.

Once the paint is dry, mount the hardware. Use machine screws and nuts and do not forget the star-type lockwashers. (The fasteners are not on the parts list – I get this kind of small mechanical stuff from a local hardware store.) Do not use sheet metal screws to mount anything other than the cover, as you will be sorry in a year or two if you do. The heat/cooling cycle will loosen up the sheet metal screws eventually. Based on my experience, in the worst cases you can only tighten sheet metal screws down a few times before they start to get sloppy and refuse to tighten up any more. Also, they leave a sharp point exposed on the inside of the chassis.

Mount the tube sockets, but leave the tubes safe in their boxes until it is time to fire the amp up. I wired the power supply first. I put two terminal strips side-by-side on one of the transformer mounting bolts. Sand down to bare metal under the feet of the terminal strips to get a good ground connection.

Make a ground bus (the bus is just a common wire for grounding everything). The middle lug of the terminal strips is used for ground. I didn’t have any fancy bus wire, so I just stripped some of the 22ga solid copper wire, and made a ground layout.

Figure 6

On the self-illuminated switch from Radio Shack, there are three connections, two of which act like a regular SPST switch. The third connector shows an open circuit with the other two no matter what position the switch is in. That is the one with the neon bulb in it: neon bulbs have infinite resistance until they reach their working voltage, and then they arc across to create light. So wire the center lug to AC line in and the regular switch lug to the transformer primary hot side. The 200K resistor connected to the third lug dims the light a bit because otherwise it is too bright.

At $35 (US), the Hammond 269AX is the single most expensive part of the project, but then again the transformers usually are in a tube project. The heater filament winding can provide 2A at 6.3VAC, and the high voltage winding is 100mA at 250VAC. Since the nominal current draw of the amplifier is 27mA, that allows plenty of reserve. Hammond makes a 369AX, which has the same specs as the 269AX but a “universal” primary winding that accommodates line voltages from 100VAC to 240VAC.

Physically locate the six large audio circuit capacitors. You want to be sure there is room for them, but put them in last, since they are very difficult to work around. The rest of the wiring is kind of like building a layer cake. The upside-down chassis is like a cake pan, and you want to build layers in the pan from the bottom up. As you wire each component, the leftover wire ends that you clip off have a tendency to fall down into the chassis. It only takes one stray wire end to cause a short circuit. With a chassis this small, it is a good habit to turn the chassis right-side-up and shake them out as you go along, right after they fall in.

Proper point-to-point wiring practice says that you should run anything with an alternating current in twisted pairs. This would include power supply wiring as well as signal wiring. I didn’t do this, as the photos show. Shame on me!

Wire each tube socket, and try to get the best fit for the resistors and small capacitors, making them physically secure and unlikely to short against each other. Then wire the input circuits. The last step is to install the six large capacitors (four 220uF electrolytics plus the two 0.47uF 400v orange drop film capacitors).

As usual with a prototype, the wiring in my amp looks a bit like a rat’s nest, since I unsoldered and resoldered almost everything, for one reason or another, before I was done. But it all fits in easily. There is plenty of room for a crossfeed filter near the input jacks if you want to put one there. (I use a Jan Meier crossfeed filter, but have it in a small project box connected between the CD player and the amp.) Once you are done wiring, look everything over systematically to make sure that all is wired properly. If you can, have someone else look it over, too. The wiring guide is handy for this.

Testing the Amplifier

Put the tubes in the tube sockets. Do not put the bottom cover on yet. Prop the amp securely on its side so you can get at the wiring inside. If you have access to a Variac, use it to slowly bring the voltage up to line level. Otherwise, the fallback method is to stand back and turn it on. Once in a while electrolytic capacitors have been known to pop, and they can spit small amounts of hot gel when they go, so make sure your hands and eyes are several feet away and out of range. If nothing hisses, pops, splutters, or explodes you are halfway there.

The best way to test the high voltage supply is actually to power up the amp with the tubes in place; otherwise the power supply voltages climb up to 400v+ with no load, which doesn’t really say much. To trim the high voltage supply, change the value(s) of the paralleled 1K, 2W resistors. These should probably be decreased to get the voltage up to around 350v. I doubt people are going to have problems with the output being much higher than that. Given the 6v tube-to-tube variation I observed, I would think anything between 340 and 360 volts should be fine. When I run the amp and watch it for 30 minutes or so, the high voltage supply drifts slowly up or down a volt or two, but the average is pretty close to 349 volts.

Checking the tube filament supply requires the tubes be installed in the sockets and be at operating temperature. The heaters draw about as much current as the transformer can produce, so it pulls the DC voltage down to about 6V. After the amp was assembled, I measured about 1.9 amps at 5.9 volts. Once the power supply works, wire the audio circuit. Do not forget to discharge the power supply and output capacitors first!

Remembering the safety rules about high voltages, check the voltages at the locations shown in red on the circuit diagram. My line voltage is high, measuring between 122 and 123 volts (the power substation is only a few hundred yards away). With the 6N1Ps in the circuit, I measured 172v at the junction of R2 and V1; 346v at the junction of R4 and V2; 175v at the positive side of the output capacitor; and 1.0v at the cathode of V1. The voltages should be within a few volts of these measurements, and the channels should be within a few volts of each other. If these check OK, shut things down, discharge the capacitors, give it one last visual inspection, and put the bottom cover on. Set it on its feet, and away you go!

Troubleshooting

I’m not good enough at troubleshooting myself to be of much help to others, but here’s my general approach.

1. Something is wired wrong!!??
2. Missing or cold solder joint? It’s easy to skip one when soldering a number of joints. Cold solder joints usually look dull instead of nice and shiny. They are easier to get when doing point-to-point wiring than when wiring a pc board – you need to be sure that nothing moves until the solder is set. Cold solder joints sometimes don’t conduct electricity at all, or they cause noise or have high resistance. If in doubt of a connection, re-solder it.
3. Really, something is wired wrong!!
4. Defective part? Not very likely since tube stuff is pretty rugged, but I suspect capacitors first. Use new-manufacture tubes at first until everything is OK for sure. Then if you get a bum NOS tube you will be able to tell that it’s the tube.
5. Don’t be embarrassed to ask for help on the Headwize DIY forum – that’s what it’s there for!

The Results

I am not a “golden ear”. I am a musician and have been an audiophile since I was quite young. I don’t believe in detailed “audio memory” that lasts more than a minute or two, so need repeated side-by-side comparisons to evaluate things. My primary headphones are the Sennheiser HD565 (150 ohms and 97dB/mW), AKG K501 (120 ohms and 94dB/mW) and Sony V6 (64 ohms and 106db/mW). My basis for comparison is the CMoy/Hansen amp (the single 9V battery version), which I think sounds pretty good, and the headphone jack of the NAD 314 (no crossfeed). I have a separate crossfeed I can use with the 6N1P amp – it’s one of the Jan Meier types, but I don’t remember which curve I followed.

Being a bass player and a drummer, I chose several of my favorite bass-player albums to use as test discs with my Denon DCM-370: “Outbound” HDCD by the Flecktones; Matthew Garrison’s self-named CD, “Bent” by Gary Willis. The last also features monster drummer Dennis Chambers. The big difference between the earlier version of the 6N1P amp and this revised version is that the new version is capable of ear-splitting volumes driving the Sonys and the Senns, and also gets very loud with the K501s. The older version worked but only at polite volume levels.

There are much bigger differences between the three headphones than there are between the three amps. All three amps make the Sonys sound like Sonys and so forth. I didn’t notice a lot of difference in sound field and imaging between the three amps, taking crossfeed into account as best I could. The CMoy/Hansen amp is a good standard of comparison, since lots of people have heard it by now. It has a typical solid-state sound: punchy and a little raspy in the upper midrange. At high volumes, the bass gets a little thin – it’s still loud, but it loses some of its rounded character.

Whatever NAD put in the 314 headphone jack, it sounds surprisingly good. It has better octave-to-octave balance than the cmoy/Hansen, eg. a more even frequency response. At loud volumes the bass is smoother and rounder than the cmoy amp. The NAD (obviously) doesn’t have crossfeed, though.

The 6N1P OTL stays smooth at high volumes, the treble is a bit more laid back than the solid state amps, but just a bit. The tube amp is just as punchy (a benefit of OTL) and the bass is very full and round, especially when using crossfeed. The amp has that tube “sweetness”, but I have to say that the NAD sounds almost as “sweet”, meaning a lack of harshness or “grain” in the sound. With the Meier crossfeed connected to the 6N1P OTL, the Senn HD565s have almost too much bass for my tastes, but without crossfeed they are just about right. Conversely, the AKGs sound a little thin without the crossfeed, but just right with it. For long listening sessions, I think the 6N1P amp, with crossfeed, with the K501s, is a very good sounding and low-ear-fatigue setup.

Overall, I can’t really identify a “tube sound” with the 6N1P amp. It just sounds very good and very smooth. I am happy with it, happy enough that I have stopped longing for a directly heated triode output transformer-less headphone amp.

Appendix 1: Simulating the 6N1P OTL Amplifier

Editor: This section discusses how to use OrCAD Lite circuit simulation software to simulate the optimized 6N1P OTL amplifier. OrCAD Lite is free and the CD can be ordered from Cadence Systems. At the time of this writing, OrCAD Lite 9.2 is the latest version. OrCAD Lite 9.1 can be downloaded from the Cadence website (a very large download at over 20M) and should work as well. There are 4 programs in OrCAD suite: Capture, Capture CIS, PSpice and Layout. The minimum installation to run the amplifier simulations is Capture (the schematic drawing program) and PSpice (the circuit simulation program).

Download Simulation Files for the optimized 6N1P OTL Amplifier

Download OrCAD Triode Simulation Libraries

After downloading 6n1potl_sim.zip and orcad_triodes.zip, create a project directory and unzip the contents of the 6n1potl_sim.zip archive into that directory. Then extract the contents of the orcad_triodes.zip archive into the \OrcadLite\Capture\Library\PSpice directory. The files triode.olb and triode.lib are libraries containing simulation models for several popular types of triode vacuum tubes, including the ones used in this amplifier. They are based on tube SPICE models found at Norman Koren’s Vacuum Tube Audio Page and Duncan’s Amp Pages.Note: heater connections are not required for any of the triode models.

The two basic types of simulation included are frequency response (AC sweep) and time domain. The time domain analysis shows the shape of the output waveform and can be used to determine the amplifier’s harmonic distortion. They both run from the same schematic, but the input sources are different. For the frequency response simulation, the audio input is a VAC (AC voltage source). The time domain simulation requires a VSIN (sine wave generator) input. Before running a simulation, make sure that the correct AC source is connected to the amp’s input on the schematic.

The following instructions for using the simulation files are not a complete tutorial for OrCAD. The OrCAD HELP files and online manuals include tutorials for those who want to learn more about OrCAD.

Frequency Response (AC Sweep) Analysis

1. Run OrCAD Capture and open the project file “6n1potl.opj”.
2. In the Project Manager window, expand the “PSPICE Resources|Simulation Profiles” folder. Right click on “Schematic1-freq_resp” and select “Make Active.”
3. In the Project Manager window, expand the “Design Resources|.\6n1potl.dsn|SCHEMATIC1” folder and double click on “PAGE1”.
4. On the schematic, make sure that the input of the amp is connected to the V3 AC voltage source. If it is connected to V2, drag the connection to V3. By default, V3 is set to 0.5V. (Note: the tubes in the OrCAD schematic are labelled U1, U2 and U3. In the article schematics, they are referred to as V1, V2a and V2b.)
5. To add the triode library to the Capture: click the Place Part toolbar button (). The Place Part dialog appears. Click the Add Library button. Navigate to the triode.olb file and click Open. Make sure that the analog.olb and source.olb libraries are also listed in the dialog. Click the Cancel button to close the Place Part dialog.
6. From the menu, select PSpice|Edit Simulation Profile. The Simulation Settings dialog appears. The settings should be as follows:
   Analysis Type: AC Sweep/Noise
   AC Sweep Type: Logarithmic (Decade), Start Freq = 10, End Freq = 100K, Points/Decade = 100
7. To add the triode library to PSpice: Click the “Libraries” tab. Click the Browse button and navigate to the the triode.lib file. Click the Add To Design button. If the nom.lib file is not already listed in the dialog list, add it now. Then close the Simulation Settings dialog.
8. To display the input and output frequency responses on a single graph, voltage probes must be placed on the input and output points of the schematic. Click the Voltage/Level Marker () on the toolbar and place a marker at the junction of R9 and the grid of U1. Place another marker just above R_Load at the amp’s output.
9. To run the frequency response simulation, click the Run PSpice button on the toolbar (). When the simulation finishes, the PSpice graphing window appears. The input and output curves should be in different colors with a key at the bottom of the graph.
10. The PSpice simulation has computed the bias voltages and currents in the circuit. To see the bias voltages displayed on the schematic, press the Enable Bias Voltage Display toolbar button (). To see the bias currents displayed on the schematic, press the Enable Bias Current Display toolbar button ().

Time Domain (Transient) Analysis

1. On the Capture schematic, make sure that the input of the amp is connected to the V2 sinewave source (the default values are: VAMPL=0.5, Freq. = 1K, VOFF = 0). If it is connected to V3, drag the connection to V2.
2. In the Project Manager window, expand the “PSPICE Resources|Simulation Profiles” folder. Right click on “Schematic1-transient” and select “Make Active”
3. From the menu, select PSpice|Edit Simulation Profile. The Simulation Settings dialog appears. The settings should be as follows:
   • Analysis Type: Time Domain(Transient)
   Transient Options: Run to time = 10ms, Start saving data after = 0ms, Max. step size = 0.001ms
4. To display the input and output waveforms on a single graph, voltage probes must be placed on the input and output points of the schematic. Click the Voltage/Level Marker () on the toolbar and place a marker at the junction of R9 and the grid of U1. Place another marker above R_Load at the amp’s output.
5. To run the time domain simulation, click the Run PSpice button on the toolbar (). When the simulation finishes, the PSpice graphing window appears. The input and output curves should be in different colors with a key at the bottom of the graph.
6. To determine the harmonic distortion at 1KHz (the sine wave frequency), harmonics in the output waveform must be separated out through a Fourier Transform. In the PSpice window, press the FFT toolbar button (). The PSpice graph changes to show the harmonics for the input and output waveforms. The input and output curves should be in different colors with a key at the bottom of the graph.
7. The fundamental frequency at 1KHz will have the largest spike. The other harmonics are too small to be seen at the default magnification. In the PSpice window, press the Zoom Area toolbar button () and drag a small rectangle in the lower left corner of the FFT graph. The graph now displays a magnified view of the selected area. Continue zooming in until the harmonic spikes at 2KHz, 3KHz, etc. are visible.
8. Harmonic spikes should exist for the output waveform only. The input is an ideal sine wave generator and has no distortion. To calculate total harmonic distortion, add up the spike values (voltages) at frequencies above 1KHz and divide by the voltage at 1KHz (the fundamental).

Additional Simulation Tips

• To change the value of any component on a schematic in the Capture program, double-click on the value and enter a new value at the prompt.
• To measure the grid-cathode voltage of tubes (V_gk), use the Voltage Differential Marker (). Click the Voltage Differential Marker toolbar button and touch the probe to the tip of the grid pin and then cathode pin.
  

Note: simulations only approximate the performance of a circuit. The actual performance may vary considerably from the simulation as determined by a number of factors, including the accuracy of the component models, and layout and construction techniques.

Addendum

3/16/2001: Revised power supply (figure 3) for 184 volts.

4/30/2002: Updated amplifier (figure 2) and power supply (figure 3) schematics to re-balance output stage for driving low impedance headphones. The author worked with Alex Cavalli to determine new values for R2, R4, and R5 in figure 2, using techniques based on the work of John Broskie. The high voltage supply has been increased from 184V to 350V by changing components of the filter network. The higher plate voltage is required to rebias the 6N1P to a better part of its operating range and to increase the idling current. The new higher plate voltage is the reason why the 6N1P tube is not really interchangeable with the 6DJ8/6922 in the Morgan Jones.

The heater supply has been given a floating ground to avoid exceeding the cathode-heater spec of the 6N1P.

The troubleshooting section and parts list have been revised to reflect these changes. Added section on simulating the 6N1P OTL amplifier with OrCAD Lite.

c. 2002 Bruce Bender.

The SESS Tube Headphone Amplifier.

by Andrea Ciuffoli

The single tube configuration is the longing of any audio designer, and here we could have only one tube per channel. If we have only one tube on the signal the distortion spectrum will be a perfect decay on all harmonics, so it sounds very natural. We have also no capacitor and no resistance on the signal so nothing can change the characteristic of sound. This project is a single-ended/single-stage/single-tube (SESS) amplifier using the Lundahl LL1630 output transformer, which is gapped for 10mA DC current. It can be used as the stage after a DAC chip (see Tube Headphone Amplifier with Digital Input) and before the headphones.

Figure 1

To design a SESS Tube Headphone Amplifier we need a triode with uncommon characteristics: enough voltage gain, low internal resistance and good anodic current. My first test was done with the E182CC (figure 1), but there is the limitation on the usable impedance of headphones in the range of 300 to 600 ohms for maximum performance. The output impedance of the amplifier is about 35 ohms, and I don’t accept any damping factor less than 8. (About damping factor: for some, a damping factor of 4 or less is acceptable, but only because they have not heard a better tube headphone amplifier like this one.) The tubes that could be used are E182CC or 5687, but with the 5687, the gain is less. The E182CC is auto-biased to ground through the attenuator. It needs the 1K input resistor for stability, probably because there is no shield on middle of the tube.

Figure 2

A while ago Stefano Perugini, designer of a very good DAC, contacted me to suggest using in this design the Russian tube 6H30 (figure 2). This tube has good gain, a very low internal resistance (near 800 ohms) and incredible trasconductance, so it is the final solution to get a very low output impedance to drive headphones from 150 ohms to 600 ohms. This tube also has incredibly low distortion, and the sound characteristic is very natural.

6H30 Datasheet: Page 1 Page 2

The 5842 version (figure 3) gives a more opened and detailed sound than E182CC and 6H30 versions, but the big advantage is that the gain is enough to be driven directly by any CD Player or DAC.

Figure 3
Download Raytheon 5842 Datasheet (PDF)

The LL1630 output transformer (also used in my Tube Headphone Amplifier with Digital Input) has a wide frequency response from 10Hz to 40kHz (+/-0.5dB), and a good inductance value to build a perfect headphone amplifier to drive headphones with impedance from 150 to 600 ohms. The LL1630 transformer here is used with parallel connection of the primaries. It must be gapped for a 10mA DC current, so that in parallel connection, it will run fine with the 20mA or less bias current of the output tube. When ordering this component from Lundahl, ask for LL1630/10mA.

The other components are not cheap, and here the highest quality parts must be used to get the maximum result. About passive components types, I don’t leave many choices: Holco or Caddock resistances on cathode, ELNA Cerafine or Blackgate capacitors on cathode and power supply, and the DACT stepper attenuator. NOBLE or ALPS normal attenuators don’t give the same pure sound.

Figure 4

About the power supply, I suggest this Reference Power Supply (figure 4) that gives the best result, but is not very cheap to implement. The 100 ohm resistors across the filament supply are center grounded for low noise. The rectification comes from a Hybrid Graetz Bridge (from a Fulvio Chiappetta article in the Italian magazine “Construire HiFi”) made with 1N5408 diodes and a RCA 5R4 rectifier tube. Although it has semiconductors, the sound depends on the rectifier tube, and the switching characteristic is that of the lower speed device – the tube. The hybrid tube bridge has the advantage that it can be driven from a single transformer secondary (the tube rectifier normally needs a dual secondary transformer).

To order the transformers contact Lundahl

• Unit price for LL1630 is $86.22 (US) or EURO 73.62
• Unit price for LL1649 is US $117.02
• Unit price for LL1673 is US $59.96

and to buy the 6H30 incredible tube (about $60), contact Stefano Perugini.

Addendum

2/15/00: Revised figure 2 – changed 1.5K resistor to 820 ohms. Also added datasheets for 6H30.

5/1/00: Added figure 3.

11/18/2001: The LL1638/10H in figure 4 was replaced with LL1673/20H, because Lundahl revised the specs of the LL1638.

c. 2000 Andrea Ciuffoli.
For commercial use of the circuits in this article, please contact Andrea Ciuffoli.
The author’s website: Andrea Ciuffoli’s Home Page.

Hi-End Transformer-Coupled Headphone Amplifier With Digital Input.

by Andrea Ciuffoli

About one year ago I designed the Top-Level OTL Tube Headphone Amplifier and more recently, I designed the No-Compromise Tube Headphone Amplifier, where each channel uses an E182CC tube (both sections) to match the low impedance of headphones. The results of these OTLs are very good and the sound is better than any other OTL headphone amplifier using regulator tubes such as 6080 or 6AS7.

But, in my continuing search for the best sound, I have created these projects. Looking into the online datasheets of the Lundahl web site, I discovered the LL1630 transformer. The LL1630 has a wide frequency response from 10Hz to 40kHz (+/-0.5dB) and a good inductance value to build a perfect headphone amplifier to drive headphones with impedances from 150 to 600 ohms.

There are 2 ways to better the performance of the OTL amplifers: use octal tubes (more linear than noval), or use a single stage instead two. After many listening tests and simulations on Spice3f4 to obtain improved sound and the same or better electric performance of the OTL solutions, I have designed an amplifier based on the LL1630. Later, I will show a DAC input stage using the Lundahl LL1566 and LL1527XL transformers.

The Amplifier Designs

The design is a two-stage, but with an octal tube – the more linear indirect-heating 6SN7. These tubes and the cathode follower configuration of second stage give very low distortion, low output impedance and an incredible low frequency response.

Ftl = less than 10Hz
Rout = less than 19 ohms
THD = always less than 1%

To get more gain, a 6SL7 tube could be used in input stage, but a higher power supply voltage, about 450V, is suggested. Both the 6SN7 and 6SL7 can work at 450V, requiring only changes to the bias resistors on cathode. With the 6SL7, the anodic resistance must be about 100K 1W instead 22K 2W.

About passive components types, I don’t leave many choices: Allen Bradley resistors on anode, Holco or Caddock resistances on cathode, ELNA Cerafine capacitors on cathode and power supply, and the DACT stepper attenuator. For all my new projects, I am using the very good ELNA Cerafine capacitors that I love and which can be find for a good price at Welborne Labs. The ELNA Cerafine capacitors contain super fine ceramic particles which, through chemical reaction, improve the discharging speed between the anode and electrolyte with very low distortion. Any other electrolytic or polypropylene capacitor is trash. Only the Blackgate WKZ could give better sound, but I have not tested it.

For interstage capacitors, the best choice would be copper film-paper in oil, but now I am using the ERO KP1832 by SteinMusic and the sound quality is very high.

I am using a DACT stepper attenuator instead of the normal ALPS or NOBLE solution, because the the sound is clearer. A good rewiew of these components can be found on the article published in Hi-Fi World. At this time I am using this amplifier with the Sennheiser HD580 (300 ohms).

The LL1630 transformer here is used with parallel connection of the primaries. It must be gapped for a 5mA DC current, so that in parallel connection, it will run fine with the 9.5-10mA bias current of the output tube. When ordering this component from Lundahl, ask for LL1630/5mA. The maximum current output is 10ma (bias current of output stage) x 7.2 (transformer ratio) = 72mA.

About the power supply, I suggest this above design that gives the best result, but is not very cheap to implement. The GZ34 Mullard could be replaced with 5R4 RCA or mercury type tube diodes.

To reduce the cost or to make a first test of this amplifier, you can build the following simple but good power supply using an isolation transformer (220V:220V or 110V:220V) from Plitron.The capacitors could be motor-start MKP types. The inductors could be neon-type inductors for 65 Watts, but, of course, the neon inductors have no gap, so some distortion is generated by the DC current flow.

Here are some test results for this project:

low frequency cut-off is about 13Hz, but here I am using the LL1630/10mA instead LL1630/5mA.

Only second and third harmonics are present up -60dB. The decay is linear, so is very natural like the other single-ended designs. In any case, what is very important is that the sound is more natural than the OTL versions!

The Digital Input Stage

About the DAC input stage, I have many solutions:

1) The Extreme DAC board by DiyZone (output stage wth op-amp is not used). Email: sales@diyzone.net.

2) The Stefano Perugini board, available for $30, including a very good power supply using a tube diode rectifier for the analog supply of the DAC.

3) the Evaluation Board CDB4390 from Crystal Semiconductor (output stage wth op-amp. is not used).

All these DACs use the same chip set: CS8412 and CS4390. The above schematic shows the basic modifications. I suggest putting the LL1566 Pulse transformer at the input in any DAC solution when not using an optical cable. The best design would use 2 LL1566 transformers, as specified in the application guide on the LL1566 datasheet. The LL1566 serves to separate the transport from the DAC.

I am using now the Crystal CDB4390 evaluation board (about $150-$180 – in Italy 340.000lire), because it comes pre-assembled. To get the Crystal CDB4390, check the Sales Offices, Representatives, and Distributors on the Cirrus Contacts page and for Italy contact:

ELCOMI
via Cassanese 27
20090 Segrate (MI)
email: elcomia@tin.it

I will start soon to test the new Crystal CDB4397, which uses the new Crystal DAC chip CS4397, the first audio DAC that offers both 120 dB dynamic range performance — the highest in the industry — and compatibility with the next generation audio formats: DVD-Audio (24-bit, 192 kHz sampling rate), SACD (Direct Stream Digital Technology).

The above diagram shows how to connect the LL1527XL to the Crystal DAC. The output stage active filter is replaced by a Lundahl LL1527XL transformer, which is configured as a bandpass filter. The RC network at the input of the LL1527XL is a high pass filter. It has a threshold frequency ft = 1/ (R * 2 * pi * C). With values of 10K and 22uF, the network has a threshold frequency of about 0.72Hz (-3db).

The high frequency cutoff is the high frequency limit of the LL1527XL (about 150kHz – see the LL1527XL datasheet). The DAC output noise is very low until 128xFs, where we can get the clock. The LL1527XL output stage does not have the same high frequency cutoff as the opamp filter (about 50kHz @ 12dB/octave), but a wider high frequency range. The 2-pole active filter used by Crystal (and also by Burr-Brown) gets the lower distortion value to show on the datasheet, but I don’t search the lowest distortion, but the best sound. The high frequency rolloff is only 6dB per octave; I cannot detect any digital noise at the output of the LL1527XL, so this is enough to use it.

This modification will only work with DACs that have push-pull outputs. With single-ended DACs, such as the Burr-Brown DAC, a capacitor should be added before the transformer to reduce the high frequency noise of the DAC. This circuit should be tested to prevent problems.

The DAC sounds better (very natural) without the active filter, because with the LL1527XL, these is no feedback on all the system, and so no compression of the sound. The difference in sound with the active filter is terrible (another world) !!!! Then you need only to assemble a simple power supply such as the following:

To order the transformers contact Lundahl

Unit price for LL1630 is US$ 86.22 or EURO 73.62
Unit price for LL1527XL when bought directly from Lundahl
using credit card payment is EURO 53.65 (VAT included)

Addendum

1/4/2000: updated DAC section.

c. 1999, 2000 Andrea Ciuffoli.
For commercial use of the circuits in this article, please contact Andrea Ciuffoli.
The author’s website: Andrea Ciuffoli’s Home Page.

A Single-Ended OTL Amplifier for Dynamic Headphones.

by Aren van Waarde

For many years, I have used a solid-state headphone amplifier in my shack. It is an op-amp type of circuit which is built with discrete components (BC559 and BC560 transistors, with a BD139/BD140 output pair). It runs in class A and has no output coupling capacitor. The measured specifications are quite good and it sounds decent after it has warmed up for about 20 minutes. However, I started wondering how a tube circuit might sound. In the stereo of my living room, I now use a tube preamp (Curcio Daniel) which sounds excellent, both on CD and MC phono.

Rudy van Stratum has published a schematic for a tube headphone amplifier in the April and September 1995 issues of the Dutch magazine Audio & Techniek. It was just a circuit idea, without any guarantee that it would actually work. The schematic caught my attention for the following reasons:

• Extreme simplicity (only 2 double triodes required for stereo). This is the simplest tube headphone amplifier that I have ever seen!
• Capable of driving low impedance headphones
• Amplifier stages directly coupled
• No global negative feedback
• Single-ended topology

I decided to give the circuit a try, and after prolonged listening tests (more than 3 months, with both CD and analog tape as signal sources) I can report that it works very well.

THE AMPLIFIER

Figure 1

My own (slightly modified) schematic of the headphone amp is shown in Fig. 1. The first stage uses one-half of a E88CC (6922/6DJ8/ECC88) in a common cathode configuration. This is directly coupled to a cathode follower which employs one-half of a 6AS7G. I have added a volume potentiometer and a grid stopper resistor to the original schematic. The size of the output coupling capacitor was also increased (from 100 to 220 uF), simply because I had this value in stock and also because I intended to use 32 and 60 Ohm headphones. With 60 Ohm headphones, the calculated -3 dB cutoff point is now at 12 Hz, whereas with 32 Ohm phones, it is at 22 Hz.

A prototype which I made on a piece of plywood worked immediately and I really liked it. With good recordings there is a life-like quality to the sound. Voices and instruments are pinpointed on the stage, with lots of musical detail and “air”. Cathode followers have the reputation of “muffled” and “boring” sound, but Rudy’s circuit renders dynamic contrasts very well and it grips your attention. Minor details in recordings become audible. One can hear, for example, the difference between different violoncellos, and the fact that different tracks of a CD are recorded in a slightly different recording venue.The solid-state amp sounds “hard”, somewhat “glassy” and “mechanical” in comparison, with less detail and less precise imaging. This surprised me greatly since the tube amp has an output coupling capacitor which the solid-state amp lacks. Apparently, the absence of global feedback and the simplicity of the tube circuit works wonders for the sound. The single-ended topology of the tube circuit may also result in a spectrum of harmonics which is different from that of the solid-state amp which has a push-pull topology.

Since I am quite happy with the sound, I have now built a final version of the circuit on a matt black aluminum chassis (size 4 x 8 x 1 inches). The amp is hard-wired, without any PCB. For my listening tests, I used Sennheiser HD 465 (60 Ohms) and Panasonic EAH-S30 headphones (32 Ohms). However, I suspect that the tube amp would sound even better when 600 Ohm headphones were used such as the Sennheiser HD 580 or the AKG K240.

THE POWER SUPPLY

Figure 2

Modifications of the power supply have a marked effect on the sound quality of this simple amplifier. For the initial listening tests, the amp was fed from the high-voltage power supply on my workbench. Then I tried it with a regulated solid-state power supply (schematic with 2 x BF459 from a book by R. zur Linde). The regulated power supply was not an improvement (as I had expected) but it caused a sonic degradation! The “magical quality” was gone, the amp began to sound like its solid-state counterpart… Subsequently I wanted to try a power supply with a vacuum tube rectifier, but the EZ81 which I intended to use proved hard to obtain. Finally I settled for an inductor-smoothed power supply with solid-state rectification (Fig.2). This simple circuit sounds very well.

The tube heaters are fed from a DC supply with a LT1084CP regulator (see Fig. 2). Since this IC dissipates about 10 W of power, it is bolted to the aluminum chassis. The rectifier diodes get hot too, and they are mounted at some distance of the chassis, with sufficient ventilation. The 1k variable resistor (P2) is used to adjust the output voltage (6.3 Volts, loaded).

I have not included details for the primary circuit of the transformers in Fig. 2. Please use the appropriate fuses for your transformers and mains voltage. My power supply has a mains switch (which activates the heater power supply) and a standby switch (which brings +150V to the plates after about 30 seconds of heating). The power supply is hard-wired and built on a separate chassis (12 x 6 x 2 inches).

MEASUREMENTS

Since I am a hobbyist, I have limited possibilities for amplifier measurements. Here is the only data I was able to collect:

Frequency response (-1 dB):
< 10 Hz…. > 100 kHz (0.775 V out in both 60 and 600W) (I therefore suspect that the output capacitors have a larger value than the specified 220 mF. NB My sinewave generator runs only from 10 Hz…100 kHz)

Output voltage:
ca. 28 Vtt in 600 Ohms (onset of clipping) = 10 Veff
ca. 3.7 Vtt in 60 Ohms (onset of clipping) = 1.3 Veff

Max. power output:
170 mW in 600 W
28 mW in 60 W

Voltage gain:
8 x (i.e. 100 mV at the input produces 800 mV output at a 600 Ohm load, volume potentiometer at maximum)

Squarewaves (1 kHz, 10 kHz, 20 kHz) look perfect, at low and very high frequencies (smaller than 100 Hz or greater than 50 kHz) one sees the influence of the output coupling capacitor.

I think these specifications are good, but the best measuring instruments are the human ears.

PARTS LIST (AMPLIFIER)

P1 – Potentiometer 100 k logarithmic stereo (ALPS RK-27112)
R1 – 1M ohms, 1 Watt carbon resistor
R2 – 33 ohms, 0.5 Watt metal film resistor
R3 – 47K ohms, 1 Watt carbon resistor
R4 – 820 ohms, 1 Watt carbon resistor
R5 – 4k7 ohms, 5 Watt wire-wound resistor
R6 – 3k3 ohms, 10 Watt wire-wound resistor
R7 – 10k ohms, 0.5 Watt carbon resistor

C1,C2 – 220 uF, 400 V electrolytic capacitor (Nichicon)
C3 – 220 uF, 100 V electrolytic capacitor (Nichicon)
C4 – 0.22 uF, 250 V MKT (DDR stock)

V1 – E88CC (Brimar)
V2 – 6AS7G (RCA)

1 – noval chassis-type tube socket (ceramic, gold-plated contacts)
1 – octal chassis-type tube socket (I used an octal socket for an Omron relay !)
2 – RCA jacks (gold-plated, insulated)
1 – 6.3 mm stereo jack for headphone plug
1 – knob for the volume potentiometer
1 – aluminum cabinet 4 x 8 x 1 inch (black, Monacor, for Eurocard PCB)
1 meter Prefer microphone cable (for wiring)
1 meter wire red
1 meter wire black

Please note: C1, R5 and C2 are shared by both channels.

The shield between the two halves of the E88CC is grounded.

The heater power supply doesn’t float, but is grounded to avoid the pickup of hum.

With shorted inputs, or a low impedance signal source, the amplifier is completely free of hum and noise, even at full volume. In practice, the volume control is never increased more than half-way.

PARTS LIST (POWER SUPPLY):

P2 – 1k trimpot (Piher)
R8,R9 – 6.8 Ohm, 1 Watt carbon resistor
R10,R11 – 180 Ohm, 0.25 Watt metal film resistor

C5,C6 – 22 nF, 1 kV MKT (DDR stock)
C7,C8 – 100 uF, 450V (F & T)
C9 – 1 uF, 250 V MKT (Philips)
C10 – 22000 uF, 25 V (Sprague Powerlytic)
C11 – 10 uF, 63 V (Philips)
C12 – 100 uF, 35 V (Roederstein)

IC1 – LT1084CP (Linear Technology)
D1,D2 – 1N4007
D3..D6 – P600A (50V, 6A)

T1 – 220:2 x 115 V, 30 VA isolation transformer
T2 – 220:9 V, 50 VA transformer
L1 – Inductor 10 H, 90 mA, 270 Ohm (Triad)

c. 1999, 2001 Aren van Waarde.
The author’s website: Aren’s Attic.

Addendum

5/28/01: Mike Fieger built the van Waarde amplifier on two chassis, one for the amp and the other for the power supply for performance and space reasons. He writes: I like to see only the amp part sitting on my table, not a big box. The fan noise is not a problem, since the power supply is far away from my ears. This amp seems to be quite exceptional.

I tested the amp by using function generators, and oscilloscope, and a multimeter. Dr. Aren van Waarde has done a great job in objectively measuring the amp’s performance. Results were quite close to those published in the article. The last 80 percent of volume usually resulted in clipping, easily seen on an oscilloscope, on all frequencies. This was true for what ever load was being tested (30 ohms, 60, 150, 300, 600). On the higher loads, the volume would be so high, it would easily make a person deaf. The higher the load, the higher the ouput power. The amp seems to like high impeadance headphones.

Test Equipment

SONY MDR-CD360 – headphones
SONY Receiver STR-D911 – headphone jack
Sennheiser 580 – headphones
Nakamichi Receiver 2 – headphone jack
Pioneer DEH-P600 – cd player
Kenwood DP-97 – cd player
Heathkit IP-32 – power supply
BK Precision model 2120B – 20Mhz oscilloscope
XR2206 – Function generator
load resistors: 30, 60, 150, 300, 600 ohm

With my cheap (32? ohm) headphones, the sound was quite bad. With these headphones (Sony MDR-CD360) full volume on the amp was extremely distorted, and not that loud. However, I borrowed a pair of Sennheiser HD-580’s, and those made this amp sound really good. Sounds very musical, and even half way on the volume was very loud. This was my second tube amp project, and I’m quite impressed with it. The Sennheiser 580-6AS7G combo sounded beter than the 580’s in the other amp’s headphone jack. Quite better. The sound was clearer, and more open. Not muddy, as in the other amp’s headphone jacks. The 6AS7G is by far the most enjoyable source of music used for listening tests. No buzz was heard when using the described power supply, or when using the Heathkit IP-32 power supply.

I constructed the amp using point to point parts placement. With many resistors in series, parallel, or both, this resulted in quite an ugly mess. I used multiple ground points, and just sort of randomly placed parts. No predetermined parts placement was used (hence the mess). I found no mistakes in the circuit during construction. This amp is very easy to build, should be perfect for the novice (like me).

With some exceptions, all of the resistors and capacitors can be ordered from Digikey. All part numbers used in builing the amp have been included in the list. As seen in the pictures, most of the resistors had to be in parallel, series, and or both. Unless the proper value resistors can be found, building the circuit on a circuit board would be highly recommended. IC1, the LT1084CP voltage regulator proved to be difficult to find, so I substituted a NTE970. No circuit modifications were necessary; the NTE970 just dropped right in. The NTE970 must be mounted on a heatsink, and since it gets very hot, I also used a CPU fan.

Download Mike Fieger’s parts list for the Waarde amplifier

The power supply requires the isolation transformer as described in the article. I did a fairly poor joob of choosing parts, and placing them into the amp. This means that the amp can only sound better with better construction techniques. Of course that means using proper and or better quality components, and good construction practices.

12/17/2001: Ying Mingyao wrote: I have finished my first headphone amp according to the single-ended OTL amplifier from Aren van Waarde. The amp is easy to build and it has a true tube sound and very excellent performance. In order to have a true tube sound, I changed some parts of Waarde’s original design, especially in the power supply.

I use two 6Z4 tube recifiers (a Chinese tube, same as 6X4) and a choke. The tube filament supply of 6AS7G is AC instead of DC – there isn’t any noise also. V1 is either a E88CC or 6DJ8. V2 can be a 6AS7G or 6N5P or 6080. The 6AS7G tube is not easy to find here, so I use the Chinese tube 6N5P or the USA 6080 tube; they all give good results. The 6DJ8 filament supply uses a 7806 regulator. In my amp, the 7806 IC doesn’t get hot. If there is room in the chassis, a small heatsink can be used but without it the IC will also be OK and very safe. The 6AS7G filament uses 2.5A. It is not regulated, because there will more heat and power wasted.

My headphones are the AKG K501 and an Aiwa 36-ohm headphone. They all perform well with this tube amp. I also tried a 16-ohms headphone and the result is good. So I can say this tube amp can really drive low impedance headphone very well.

4/14/2002: Kaj Toivola built this Waarde headphone amplifier with dual mono volume controls for the left and right channels (DACT 100K stepped attenuators). The two switches on the left are for switching the heater supply and the high voltage supply separately. Toivola owns many headphones such as the Grado SR80 and SR325, the Sennheiser HD580/HD600/HD265 and the Koss PortaPro. He mostly uses the Grados and Sennheiser HD600 and says that the amp works better with the HD600 than the Grados.

4/18/2002: Alan Buckbee‘s Waarde amplifier (shown below) has a modified power supply, because the original supply ran too hot. He also supplied a list of US-based sources for the amplifier components. He writes:

The Waarde amp is a great design. It is dead quiet, no hum, etc. Vocals (eg. Diana Krall) have such a rich texture, it is hard to believe that it is a CD! It is as if she is in the same room. Piano is incredible. One can plainly hear the decay in the note just as if one was in the same room with the piano. The air around the music is spellbinding. For headphones I primarily use the Sennheiser HD600.

Mr. Aren van Waarde built his amp using parts primarily obtained in Europe. I built this amp sourcing parts from the USA (see list below). As a result I had to make modifications from the original design to accommodate USA-sourced transformers using standard USA voltage. I built the amp in two separate boxes.

The Headphone Amplifier:

Not many modifications to the original amplifier design except that R6 is a 25 watt, wirewound resistor. R6 gets extremely hot using a 10 watt resistor. So to dissipate the heat better, I went to a 25 watt resistor. For capacitor C4, I have experimented with .22uF 600V “Orange Drop” 716Ps and some Angela Fast Caps. No opinion as to which is better at this time. I also used an Alps Blue dual/stereo 100k potentiometer for the volume control.

Power Supply Section:

This section required that I make many changes. The filament supply (6.3 VDC) line got extremely hot. It seems that both D3..D6 and the IC1 voltage regulator ran way to hot for my liking. The initial T2 transformer (Hammond 167N6) 4 amp supply did not have enough headroom to drive both the 6AS7G and the 6922 without getting extremely hot (the 6AS7G draws about 2.45 amps and the 6922 draws about 55 mA). After 1 or 2 minutes, I could not even touch the voltage regulator or the bridge rectifier.

Within 5 minutes, the transformer was very hot to the touch as well. So I split off the 6AS7G from the 6922 tube, and put the 6AS7G on the T2 transformer. I ran the 6922 off of the 6.3 volt 2 amp supply built in to the Hammond 369AX transformer. The new T2 transformer (Hammond 167S6) is rated at 6.3 volts 10 amps. This change along with using a higher rated bridge rectifier (35A vs. 12A) removed most of the heat from the filament line. The 35 amp bridge rectifier is attached to the chassis to aid as a heat sink. The new T2 transformer now runs at a cool 40 degrees C. I had to add R15 and R16 to get the voltage down from 6.8VDC (based on a line voltage of 124 VAC) to 6. 1 VDC at the socket pin. I ran the 6AS7G with as low as a 118 VAC input (using my variable transformer). At this setting, the filament voltage was 5.76 VDC at the tube pin. The headphone sounded just fine even at this lower voltage. Currently I am running the high voltage line at 150 to 151VDC, the 6922 at 6.01 VDC (measured at the socket pin), and the 6AS7G at 6.027 VDC.

I also added R12 to drop the output voltage down to the 150 to 151VDC range. Without this resistor, the high voltage was running at 158 VDC.

I also added the SW1 and SW2 (and their associated LEDs) to the design. It is preferable for longer tube life to apply the filament voltage (SW1) first (for 20 to 30 seconds) prior to turning on the high voltage line (SW2). When powering down turn off the SW2 first then SW1 switch.

For those who are new to tube designs it is absolutely critical that after transformers T1 and T2 have been attached to the chassis, one place the choke L1 on the chassis and either hook up a set of headphones or a volt meter and experiment with the position of the choke on the chassis. The windings of the transformer will interact with the choke windings, causing very noticeable hum. However, with judicious placement of the choke, it is possible to place the choke on a 12″ x 8″ chassis and get an absolutely dead quiet background.

I ended up placing the choke towards the rear of the power supply and at an angle. This was determined by using DVM hooked up to the choke prior to placing it in the circuit. For this test, the choke is NOT connected to the power supply but is placed on the chassis near the transformers. Any induced voltage from the T1 and T2 will still influence the coil in L1. After turning the supply on, I moved the choke around to find a position of least voltage across the coils. The voltage across the choke is extremely low; however the noise caused by T1 and T2 is very apparent. I then hooked up my headphones to the L1 (in place of the DVM) and noted that the position selected was absolutely dead quiet! This amplifier is totally quiet even at full volume! I spent the few extra dollars to get the enclosed choke Hammond 193B verses the open-frame 158M. An enclosed choke provides better shielding and it looks better (matches T1 and T2).

Optional items were the connectors between the two chasses. One could hard wire them together or use a connector. I chose an AMP series 97 connector (unfortunately, they are expensive) as this makes it easier to move the chasses around and to place the power supply well away from the headphone amplifier. There are a few other connectors that are less costly but most are not designed to carry 150+ volts or 2.5 amps.

On my amp, the cord connecting the power supply is five feet long. This allows me to place the power supply out of sight, if so desired. I have compensated for this length by adjusting the voltages to be about 6.01 VDC at the pin for the 6922, 6.027 VDC for the 6AS7G, and 150 VDC for the high voltage line. These are all based on a 123 VAC input voltage.

If anyone that has built this amp and has some suggestions I will be more than happy to hear from you.

The following parts are the ones I used and sourced in the USA.

Waarde Power Supply Parts List

R14 (size accordingly to match brightness of R13)

P2	1.0K OHM ¾ watt ceramic pot	3009P-102-ND	Digi-Key
R8, R9	6.8 Ohm 1 watt 5% metal oxide resistor	6.8W-1-ND	Digi-Key
R10, R11	180 ohm 1 watt 5% metal oxide resistor	180W-1-ND	Digi-Key
R12	75 ohm 2 watt 5% metal oxide resistor	75W-1-ND	Digi-Key
R13	1.2 K ohm 1 watt 5% metal oxide resistor	1.2KW-1-ND	Digi-Key
R15, R16	.36 ohm 10 watt wirewound 5% resistor	900-1015	Radioshack.com
C5, C6	.022 uF 1250 v metal polypropylene cap	P10486-ND	Digi-Key
C7, C8	100 uF 450 V Elect. Panasonic TSHB cap	P10155-ND	Digi-Key
C9	1.0 uF 250V metal polypropylene cap	PF2105-ND	Digi-Key
C10, C13	22000 uF 25V Elect. Panasonic TSHA	P6591-ND	Digi-Key
C11, C14	10uF 63 V Aluminum Elect. M series	P5189-ND	Digi-Key
C12	100 uF 35 V Elect . FC series	P10294-ND	Digi-Key
IC1	Adjustable 5A Low dropout Voltage reg.	LT1084CP-ND	Digi-Key
D1, D2	Diodes – 1 A 1000V DO-41	1N4007GICT-ND	Digi-Key
D3..D6	Bridge rectifier 12A 50V GBPC	GBPC120005-ND	Digi-Key
D&..D10	Bridge rectifier 35A 100V GBPC	GBPC3501GI-ND	Digi-Key
T1	Transformer Hammond 369AX	Hammond 369AX	Angela Instruments
T2	Transformer Hammond 167S6	Hammond 167S6	Angela Instruments
L1	Enclosed D.C. filter choke	Hammond 193B	Angela Instruments
	Aluminum Box – Hammond	HM267-ND	Digi-Key
	Aluminum Box cover – Hammond	HM283-ND	Digi-Key
SW1, SW2	SPST Switch	910-4719	Radio Shack
LED1, LED2	Red LED’s	276-330	Radio Shack
	LED Holders	276-079	Radio Shack

In addition to the above one will need soldier terminal strips, wire, rubber feet, 6-32 screws, self tapping metal screws, IEC male AC connector w/fuse holder, fuse, power cord, and heat shrink tubing.

Waarde Headphone Amplifier Parts List (one channel)

P1	Alps Blue dual/stereo 100K pot.		Angela Instruments
R1	1.0M ohm 1 watt 5% metal oxide resistor	1.0MW-1-ND	Digi-Key
R2	33 ohm 1 watt 5% metal oxide resistor	33W-1-ND	Digi-Key
R3	47K ohm 1 watt 5% metal oxide resistor	47KW-1-ND	Digi-Key
R4	820 ohm 1 watt 5% metal oxide resistor	820W-1-ND	Digi-Key
R5	4.7 K ohm 5 watt 5% silicone resistor	45F4K7-ND	Digi-Key
R6	3.3 K ohm 25W 5% wirewound resistor	900-1359	Radioshack.com
R7	10K ohm 1 watt 5% metal oxide resistor	10KW-1-ND	Digi-Key
C1, C2	220 uF 400 V Elect. Panasonic TSHB cap	P10145-ND	Digi-Key
C3	220 uF 100V Elect. Panasonic FC	P10780-ND	Digi-Key
C4	.22 uF 600V orange Drops 716P		Angela Instruments
V1	6922-E88CC	T-6922_E88CC	Antique Electronic Supply
V2	6AS7G – triode, Dual Svetlana	T-6AS7G-Svet	Antique Electronic Supply
	9 pin ceramic/gold chassis mount socket		Angela Instruments
	Octal ceramic/gold chassis mount socket		Angela Instruments
	RCA Jacks – Vampire	M1F/OFC	Welbourne Labs
	6.3 mm (1/4″) stereo phone jack	SC1107-ND	Digi-Key
	Knob for P1	274-0424	Radio Shack
	Aluminum Box – Hammond	HM260-ND	Digi-Key
	Aluminum Box cover – Hammond	HM277-ND	Digi-Key
Optional Items:	Amphenol 97 series threaded connectors (see text)
	Connector plug 7 position w/pins	97-3106A-16S-1P-ND	Digi-Key
	Conn. Recept. Box mnt 7 pos w/soc	97-3102A-16S-1S-ND	Digi-Key
	Cable clamp w/bushing	97-3057-1008-1-ND	Digi-Key

In addition to the above one will need soldier terminal strips, wire, rubber feet, 6-32 screws, self tapping metal screws, power cable (6 conductor) between boxes, and heat shrink tubing.

9/30/2004: James Deaver (aka vaklov in the forums) built a Waarde amp with a glass chimney over the 6AS7G to convection cool the power tube. The amp is named the “Neva.” He upgraded the 3.3K ohm cathode resistor, R6, to 20W (CADDOCK MP820 film power resistors – chassis mounted), because the original 10W got very hot, very quickly.

The power supply was split into separate low and high voltage sections as separate circuits, so that the amp could run in “standby” mode, or for “burn-in” heating of new tubes before high voltage is applied. Each section has independent, switching, fusing, and indicators. He used Hammond power transformers (model 166P10 @ 10V/5A and model 261G6 @ 250VCT/45VA) to get a B+ of 200VDC, adjustable downward (he prefers an “at plate” reading of about 170 – 175VDC on the 6AS7G).

A 6NO60 octal delay tube (Amperite) is connected in series with the HV primary for a 60-second turn-on delay to allow the tubes to stabilize before turning on the high voltage supply. The LT1084 filament supply regulator was running hot (the two filaments draw 2.8A continuous and the delay tube another 0.5A), so he substituted the LT1083, which is rated at 7.5A.

He writes: I am a disabled vet who chose to build Dr. Aren van Waarde’s OTL amp, and although it took me 2 1/2 years the result is well worth it! I am sufficiently proud of my work that I have devoted a section of my website with pictures of the units and some construction details/mods I would like to share with the other DIYers.

The Neva is named for Stephen and Neva Allen, two dear friends now living in Maine (amplifiers are always female). It uses a glass chimney surrounding a ceramic octal base to convection cool the 6AS7G power tube and the circuit components inside the enclosed chassis (via filtered/screened vent holes in bottom). The concept is VERY old and VERY simple: using a cylinder to surround a heat source to draw air in the bottom of the cylinder and forcibly eject it out the top, creating a Venturi Effect or “heat pump”. The chimney in this amp was a glass hurricane lantern chimney/globe.

About the 6NO60 delay tube in the HV supply, delay tubes were standard in the military. EVERY UNIT WITHOUT EXCEPTION that I worked on while in the U.S.A.F. used plate power delay designs! They were mostly 15- and 30- second delays switched by thermal contact relays in a tube format. Delay tubes VASTLY extend the operational life cycle of power tubes ESPECIALLY when very high plate voltages are used ( >350VDC ). In the realm of audio equipment, I believe they also minimize the annoying “turn-on THUMP” effect.

In the Neva, I connected the filaments of the delay tube with the other circuit filaments, and wired the contact pins in series with the HV transformer’s primary. I recommend “bridging” the contact pins with a small disk capacitor to minimize contact arcing. You occasionally find these on Ebay under “Time Delay Relay” or they are available from Antique Electronic Supply for about $15 US.

The second picture (top view) shows the two one-third arc openings, into the chassis interior (note the wiring for the 6AS7G socket visible in the “upper” of the two). The octal socket is top mounted for better weight loading (probably unnecessary), suspended in middle of the 2 1/2″ opening, and is not visible with tube and chimney in place.

The third picture (bottom view) shows brass screened vents to permit room air to be drawn into the chassis interior by the HOT air (155° F) exhausting from the chimney. Try to position them under the hottest circuit elements, ie. tubes, cathode load resistors, etc.

Here’s the Neva in its “dedicated” hutch. The headphone stand is made from two pieces of “scrap” American Black Walnut left over from fabricating the wood sides for the amplifier module. Add a three inch piece of “freeze proof”, black foam, water pipe insulation my plumber left behind ( with the bill! ). The vibration pad under the amplifier is two 12″ sq. ceramic tiles (surplus from a tileing job), painted flat black, with a 3″ thick block of graphite impregnated, acoustically “dead” foam, left over from unpacking an IBM 6300 Series Winchester disk drive, sandwiched in between. Result: Zero microphonics from the JAN 6922 <:-]

For more construction information, see the James A. & Kathleen C. Deaver Homepage.

 

The Morgan Jones Mini Tube Headphone Amplifier.

by Chu Moy

Back in 1999, I received an email with an attached schematic of a tube amplifier. The sender told me that it was supposed to be the schematic for a “clone” of the famous EarMax miniature headphone amplifier. Since he had no more information about the design and had not built it, I filed it away to be referred to at a later date. Months later, I saw the schematic again in a book called Valve Amplifiers (2nd ed.) by Morgan Jones. In the book, Jones described it as a reverse-engineered version of the EarMax. That is, the schematic was not of the true EarMax, but was derived from the published specifications of the EarMax (e.g., 3 tubes, the power supply voltage, the input and output impedances). Jones had created the circuit as an academic exercise, but had not actually built it. I once again put the schematic away, hoping later to find a DIYer who could give construction details.

While doing research on the internet, I came upon a reference to the Jones design in an archive for the Sound Practices mailing list. Lance Dow, who knew Morgan Jones personally, had posted the schematic in that newsgroup way back in 1996. Dow had not built the circuit either. Given all the interest in the audiophile community about the EarMax, I thought that surely someone, somewhere, must have tried to build it. I scoured the Sound Practices archives, downloading year after year of digests, and finally found a posting by Johannes Chiu, who described enthusiastically his DIY work on this design. I contacted Chiu for more information, but apparently he no longer remembers many specifics about it. This previous version of this article was a summary of the information collected from the Sound Practices archives and from the HeadWize forums about the original Morgan Jones circuit.

Since the publication of that article over a year ago, there have been several inquiries regarding the possible topology of the EarMax Pro. The Pro has basically the same output tube complement as the EarMax, but can provide higher current output into low impedance headphones. The specifications suggested that the topology of the EarMax Pro was similar to the EarMax (both having White cathode follower output stages), but the mystery remained as to how two amplifiers with similar output stages (no output transformers) could have substantially different output characteristics. Then in 2002, Alex Cavalli submitted revised Morgan Jones circuits with new parts values, based on the White cathode follower optimization techniques developed by John Broskie. The optimized Morgan Jones amplifiers (with and without feedback) can output more than 3 times the current of the original into a 32-ohm load. The amp shown at the top of this article is an optimized Morgan Jones amplifier (without feedback) built by Bryan Ngiam.

The Amplifier Designs

1. The Original Morgan Jones Amplifier

Figure 1

Figure 1 is the schematic for the original Morgan Jones amplifier. It has a grounded cathode input stage with an idling of about 3mA. The output stage is a push-pull White follower, which provides low output impedance without the need for global feedback. It idles at about 10mA and can swing ±20mA in push-pull. The output impedance is about 10 ohms (the calculated value was 6 ohms). The overall gain of the amp is about 22 (the calculated value was 28). Jones used an ECC88 input tube. The original EarMax has an ECC81 input tube, but he felt that the low anode current required for this stage would lead to noise and gain problems with the ECC81. The EarMax output tubes are ECC86. Jones used the similar ECC88, later used in the EarMax Pro’s output as well. He rated the amplifier to drive headphones from 200 to 2000 ohms.

Figure 2

In all of following voltage and Fourier analysis graphs, the red curve is the input; the green curve is the output. The top graph in figure 2 shows the output voltage waveform into a 300-ohm load (the input is a 0.15V, 1 KHz sine wave). The bottom graph is a Fourier analysis of the output waveform to determine the harmonic distortion, which turns out to be about 2%. The output current (not shown) is 9mA, so the amplifier is driving the load with 36mW (the maximum power into 300 ohms is about 120mW).

Figure 3

Figure 3 shows the same set of graphs for a 32-ohm load, with a 0.016V, 1 KHz sine wave. Like the example in figure 2, the output is being driven to a total harmonic distortion of 2%, but here, there is a distinct imbalance between the top and bottom halves of the waveforms, because the lower impedance load draws more current and is unbalancing the push-pull configuration. The amplifier is supplying about 2.6mW into 32 ohms (the maximum power into 32 ohms is about 13mW). Thus, like the EarMax, the original Morgan Jones amplifier is not truly suited to power low impedance headphones such as the Grados, despite the low output impedance.

2. Analysis of the Performance of the Original Morgan Jones Amplifier

Theoretically, this type of output stage should be able to drive low impedance loads well, because it has a very low output impedance. In his article The White Cathode Follower, TubeCad editor John Broskie investigated the poor performance of the White cathode follower when driving low impedance loads. He discovered that the voltage drop across the anode load resistor (R4) of the top tube (V2a) varies with the current flowing through the tube. If the voltage across R4 is high enough, it will overdrive the bottom tube (V2b).

Alex Cavalli viewed the problem another way: the imbalance was caused by the gain of the V2a being greater than 1:

Assume that the output of the amplifier is shorted (an AC short at the junction of the upper and lower triodes in the output stage) and ignore the fact that the tubes are in series. Under this condition, both the upper and lower triodes are operating as simple grounded cathode amplifiers, where the output of the upper section is fed directly into the lower section.

Now the gain of the first stage is about 25. The gain of the upper triode in grounded cathode mode is about 15. If a 0.01V sine wave is applied to the input, the first stage will produce about 25 x 0.01 =.25V at its plate. Thus the upper triode sees 0.25V on its grid. The upper stage in turn produces 15 x 0.25 = 3.75V, which is coupled to the grid of the lower triode.

There are two things to note here:

• 1. the grid drive to the push-pull sections is unequal, 0.25V (upper) vs. 3.75V(lower) and
  2. the bottom triode, which has a bias of about 1.75V, is being driven hard into cutoff and positive grid.

In this design the upper and lower output sections are not working together equally and so they are not producing the maximum possible current swing. Furthermore, the enormous gain feeding the lower section makes the amp extremely sensitive and sends it out of class A mode quickly.

The maximum voltage that can appear at the grid of the bottom tube is determined by the DC biasing voltage across the cathode resistor. In figure 1, for example, the bias or idling voltage across R5 is 1.7V, so the maximum peak-to-peak voltage into the grid of V2b is 1.7V. A higher grid voltage will either turn the tube off or drive the grid positive and will push the triode out of Class A operation.

Figure 5

Each graph in figure 5 shows the differential voltages between the grid and cathode (V_gk) of the output tubes V2a (blue) and V2b (magenta) in the original Morgan Jones amplifier driving a 300-ohm load. In the top graph, V_gkfor V2b is 1.66Vp-p, which is just less than the 1.7V bias voltage across R5. The V_gk for the top and bottom tubes appear symmetrical, but have unequal amplitudes. At this V_gk, the amplifier is outputting 2V into a 300-ohm load, which is approximately 13mW. This is the maximum output power of the original MJ amplifier into 300 ohms before push-pull stage leaves class A mode. The Fourier analysis indicates that the harmonic distortion at 13mW is only 0.3% (output voltage and Fourier graphs omitted).

The bottom graph shows the V_gk waveforms when the White follower stage is severely unbalanced. The V_gk for V2b exceeds 1.7Vp-p and the shapes of both waveforms are grossly distorted. Here, the original MJ amplifier is driving a 300-ohm load with an output voltage of 3.3V (about 36mW) and the harmonic distortion has risen to 2% (output voltage and Fourier graphs omitted).

After employing a similar White follower balance analysis for a 32-ohm load, the maximum output power of the original MJ amplifier into that load is actually less what was determined from figure 3: a mere 1.6mW at 1.3% distortion (0.228V output). Even for high efficiency, 32-ohm headphones like the Grados, 1.6mW is not enough to achieve clean volume levels – especially not if the music has wide dynamic range.

Broskie concluded that in order for the White follower to perform optimally (and maintain the balance in the push-pull pair), the anode load resistor R4 should be chosen so that the bottom tube receives an identical grid-to-cathode voltage signal as the top tube. In other words, V_gk for V2a should equal V_gk for V2b (the bias voltage across the cathode resistor still determines the limit of V_gk). His solution was to calculate a lower value for the anode load resistor (which he called R_a) based on the equation:

R_a = r_p/mu = 1/G_m

where G_m is the transconductance of the tube.

Alex Cavalli provided this explanation:

The way to balance the grid drives where the output is shorted (a load of zero ohms) is to ensure that the upper output section has a gain of 1. This will cause the lower triode to see exactly the same grid signal as the upper triode. According to Broskie the effective Gm of the upper stage is:

 

G_m = (mu + 1) / (r_p + R_a)

where R_a is the anode load resistor and r_p is the plate resistance of the triode. The gain of upper stage is given by:

Gain = ((mu + 1) x R_a) / (r_p + R_a)

To have a gain of 1:

((mu + 1) x R_a) = (r_p + R_a)

Then solving for R_a:

R_a = r_p/(mu) = 1/G_m

This result is the same as Broskie’s, except that he proves this result for all load impedances.

The transconductance for a 6DJ8 is 11mA/V, so R_a ~ 90 ohms.

3. The Optimized Morgan Jones Amplifier

Figure 4

Alex Cavalli’s revised Morgan Jones circuit is shown in figure 4. It is identical the original, except that the power supply’s current rating has been doubled and three resistor values in the amplifier have been changed. R2, R4, and R5 determine the balance for the White push-pull output stage. R2 determines the quiescent plate voltage on V1 which sets the grid bias on the V2a in combination with R4 and R5. These seemingly minor changes in the resistor values have a huge impact on the performance of the amp as discussed below.

He used PSpice simulations to determine the best values for R2, R4 and R5. Although Broskie determined that the optimal anode load resistor value R4 was 1/Gm (or 90 ohms for a 6DJ8), the simulations indicated that the amplifier had better output characteristics with a higher value – 150 ohms. The output stage still idles at around 10mA. These modifications resulted in better performance into both 300-ohm and 32-ohm loads.

Figure 6

The output voltage of the Cavalli-optimized amplifier in figure 6 (top graph) was chosen by monitoring the V_gk for V2b until it reached about 2.5Vp-p, the same value as the DC bias voltage across R5. At that point, the amplifier’s output voltage into 300 ohms is 5V or 83mW, a six-fold improvement over the 13mW maximum for the original Morgan Jones. The harmonic distortion at 83mW is about 1%.

Figure 7

The performance of the optimized amplifier into a 32-ohm load is as remarkable. Again, the output voltage in figure 7 was chosen by monitoring the V_gk for V2b until it reached about 2.5Vp-p. Based on the results of the White follower balance analysis, the maximum output power of the amplifier into 32 ohms is 10mW, a six-fold improvement over the 1.6mW maximum of the original MJ amplifier, although the harmonic distortion at 10mW is also higher: 2.1%.

Figure 8

The graphs in figure 8 compare the V_gk for the output tubes V2a (blue) and V2b (magenta) for the setups in figures 6 and 7 respectively. When the top graph here is contrasted with the top graph in figure 5, the V_gk waveforms in the optimized amplifier when driving a 300-ohm load have achieved virtually perfect balance. The bottom graph shows that when the optimized amp is connected to a 32-ohm load, the V_gk waveforms are not quite as balanced (the amplitude of V_gk for V2a is slightly larger than that for V2b), but are vastly more balanced than the curves in figure 5.

Figure 9

The frequency responses of the original (top graph) and optimized amplifiers (bottom graph) driving a 32-ohm load are shown in figure 9. The low frequency response of the optimized version has a more extended low-end than the original. The overall gain of the optimized amplifier is about 8, whereas the original has a gain of 19 (the graph scales do not make the differences in gain obvious, however). With the 300-ohm load, the difference in gains is far less: 23 and 19 for the original and optimized amps respectively (graphs not shown). The drop in gain with the 32-ohm load is due to the higher output impedance of the optimized amplifier, one of the tradeoffs of optimization. The original has an output impedance of about 10 ohms, but in the optimized version, the output impedance is 53 ohms – high enough to cause strong loading effects on a 32-ohm load. More than half of the amp’s output voltage is absorbed by the output impedance in this case.

4. The Optimized Morgan Jones Amplifier with Feedback

Figure 10

While the optimized amplifier can dump 10mW into 32 ohms, adding feedback improves the performance even more. The schematic in figure 10 is identical to the one in figure 4, except for the resistors R8 and R9 that form a feedback loop and the removal of the 100-ohm grid stop resistor on V1. A separate grid-stop resistor is not needed, because of the presence of R9. Cavalli selected the feedback resistor values by examining trade-offs. Too much feedback and the gain was reduced too much. Too little feedback and there was no benefit. He tried to select values that reduced the output impedance down substantially below 32 ohms and reduced distortion, while still leaving enough gain to be sensitive to conventional CD inputs. Cavalli recommends experimenting with these resistor values to get the best tradeoff for specific headphones.

The output graphs for this amplifier driving a 300-ohm load are not shown here, but they indicate that it delivers the same power (83mW) as before, but at a lower distortion: 0.5%. Feedback also lowers the output impedance to about 20 ohms and the gain to about 6 for a 300-ohm.

Figure 11

The output characteristics of the optimized MJ amplifier with feedback into 32 ohms are shown in figure 11. The amp’s voltage gain is about 4, because the output impedance, while less, is still significant compared to 32 ohms. Again, the maximum output power under a White follower balance analysis is the same as for the non-feedback version in figure 4, but the distortion is lower: 1.4%. Thus, the primary effect of feedback in this circuit is to provide cleaner output power for low and high impedance headphones.

5. Revised Power Supplies

The power supply used by Johannes Chiu (figure 1) was “bare bones” and provided modest filtering. A 19VAC wallwart directly powered the tube filaments connected in series. A step-up transformer converted the 19VAC to about 156VAC, which was then rectified and filtered with a small 220uF capacitor to output 220VDC.

Figure 12

Bryan Ngiam and Rudy van Stratum have modified the Chiu design to reduce noise and hum. Ngiam built two power supplies (figure 12). The first supply is the closest to the original. It uses an 18VAC wallwart to power the tube filaments connected in series, a different step-up transformer and a L-C pi output filter for improved noise filtering.

Figure 13

Ngiam’s second supply has a single custom power transformer with three 6.3VAC, 500mA secondaries. Like the first supply, the high-voltage primary employs a pi filter, but each 6.3VAC secondary powers the heater of one tube. Ngiam recommends twisting the filament supply wires “to reduce AC currents into the audio circuit.” With either supply, Bryan recommends removing the 1 Megohm and 100 ohm resistors at the input stage.

Figure 14

Rudy van Stratum’s designs are shown in figures 13 and 14, and incorporate power transformers that he already had in his possession: a 250VAC/50mA unit and a 6.3VAC/2A unit. In both circuits, the tube heaters are connected in parallel across the DC filament supply. Like the Ngiam supplies, the high voltage output of Stratum’s first supply uses an L-C pi filter.

For DIYers who cannot find an appropriate inductor, Stratum’s second circuit has several stages of RC filtering with two high voltage outputs: 230VDC and 210VDC. Originally, he powered the entire amplifier off the 230VDC tap. Later, he decided to increase the filtering to the input stage supply by adding a 2.2K resistor and a 100uF electrolytic capacitor, because most of the hum was coming from the input stage. The extra RC filtering also reduced the voltage to about 210VDC. The necessity of two high voltage taps can be avoided if the power transformer is replaced with one that outputs enough voltage to give 220VDC. It might also be worth trying to power the entire amplifier with 210VDC.

Construction

Johannes Chiu constructed the original Morgan Jones amplifier. He made one change: a 100-ohm grid stopper resistor to the input tube for each channel to improve stability. The EarMax uses a 19VAC, 350mA wallwart. Chiu created the simple power supply in figure 1 based on a 19VAC wallwart. The 19VAC directly powers the tube filaments connected in series, and could be rectified to DC for lowest noise. To step the 19VAC up to 220V, Chiu used a 10V filament transformer in reverse. However, Frank Nikolajsen pointed out that a 10V filament transformer would actually give a rectified voltage of 310V. Therefore, I have scaled the transformer secondary to 14V, but recommend experimentation. I have increased the wallwart’s current spec to 1A (Chiu’s wallwart had a capacity of 840mA). DIYers in a country with an AC standard different from the US standard should select the transformer accordingly. I have drawn the power supply with a small value filter capacitor (there were no instructions from Chiu about this). DIYers will probably want to increase the value.

Several DIYers have found that Chiu’s supply can introduce excessive hum in the amp. Bryan Ngiam and Rudy van Stratum designed powers supplies with superior filtering (figures 12-14). Bryan recommends that all input cables should be properly shielded and that a star ground should be employed whenever possible. If using a ground “strip” inside the chassis, Stratum suggests experimenting with the positioning of audio grounds on the strip to get the lowest hum and noise.

Alex Cavalli recommends a minimum power supply current rating of 220V @ 50mA for the optimized and feedback amplifiers, figuring 20mA peak per output section and 8mA for the input stages for about 48mA total. Otherwise there will be major power supply sag. He also recommends that the tubes be actual 6922/6JD8. The 6N1P is sometimes subsituted by vendors and is NOT a true substitute (see Bruce Bender’s 6N1P OTL headphone amplifier for a modification of the Morgan Jones design using that tube).

Chiu used a trapezoidal-shaped chassis measuring 2″ (top) x 4″ (bottom) x 6.5″ (length) x 2″ (height), which gives a volume 1.5 times larger than the chassis for the EarMax (3.75″ x 3.5″ x 4.0″). The additional space is required by the filament transformer. The EarMax probably has a smaller custom transformer. For the volume control, Chiu selected a 100K dual audio pot from Radio Shack (RS 271-1732).

The original Morgan Jones amplifier does not have enough current drive for low impedance headphones like the Grados, and Chiu did not have the schematics for the optimized versions. Instead, he experimented with a small impedance matching transformer (1.5K/60-60 from Antique Electronic Supply) for higher power transfer to his Grados. He attached the 1.5K primary to the output of the amplifier and connected the dual secondaries in parallel, which then became the output for the headphones. On the matter of selecting the impedance transformer, Chiu writes:

People think that the output transformer is crucial in the sound, and hence must be high quality and expensive. HOWEVER, the power involved here to drive the headphones is in the milliwatts. Given that the power is so low, it greatly relaxes the requirements for the transformer. The transformer I got was about the size of a nail, and looks as cheap as many transformers found on computer modems and the like. I would also argue that the specified frequency response is perhaps 100Hz-15kHz, which most people would frown at. BUT, these specs are at full power, which could be up to 1/2 watt. At 10mW, who knows what the response is. All I know is that I tried it, and one would be hard pressed to distinguish and sonic feature that was added because of the transformer. I would encourage people to try out a few transformers. Make sure the impedances are more less correct, such that you have enough current drive, while at the same time not lose too much signal level due to the step down.

The Result

Chiu compared his original Morgan Jones amplifier to a LT1010 buffer headphone driver and a tube headphone amplifier (a “monster” 6AS7G cathode follower driven by a 6DJ8 diff amp) he had built earlier. The headphones were a pair of Sennheiser HD420. He found that possibly “because of the higher gain of this circuit, it feels to have more punch and wallop…like moving from a cathode follower pre-amp to a mu-follower, or like adding a huge cap to your pre-amp power supply.”

With a borrowed pair of Grado SR60s and the impedance converter, Chiu noted “I think the transformer will give you 90% of what is there. Maybe the bass is a tad weaker, the highs are different, but still better than straight out from those personal walkman or cd players.” He tried the Grados without the impedance converter “and the transformer is way better.” Chiu’s final verdict on this project: “there is one thing I am certain nobody would deny if they listened to it, and that is: the amp is really fun to listen to.”

DIYers building the Morgan Jones amplifier today should try the optimized versions, which garner favor through lower distortion, higher output power and more stable output stages that benefit both high and low impedance headphones. Although low impedance headphones were starved for current with the original Morgan Jones design, the optimized amps can drive these types of headphones to reasonable volumes without the need for an impedance-matching transformer. Low impedance headphones not only get more power from the optimized amps, but also get a flatter, more extended low frequency response. The mystery of the EarMax Pro, at last, is solved.

Appendix: Simulating the Amplifier in OrCAD PSpice

This section discusses how to use OrCAD Lite circuit simulation software to simulate Alex Cavalli’s optimized Morgan Jones amplifier. OrCAD Lite is free and the CD can be ordered from Cadence Systems. At the time of this writing, OrCAD Lite 9.2 is the latest version. OrCAD Lite 9.1 can be downloaded from the Cadence website (a very large download at over 20M) and should work as well. There are 4 programs in OrCAD suite: Capture, Capture CIS, PSpice and Layout. The minimum installation to run the amplifier simulations is Capture (the schematic drawing program) and PSpice (the circuit simulation program).

Download Simulation Files for Alex Cavalli’s Optimized Morgan Jones Amplifier

Download OrCAD Triode Simulation Libraries

After downloading mj_sim.zip and orcad_triodes.zip, create a project directory and unzip the contents of the mj_sim.zip archive into that directory. Then extract the contents of the orcad_triodes.zip archive into the \OrcadLite\Capture\Library\PSpice directory. The files triode.olb and triode.lib are libraries containing simulation models for several popular types of triode vacuum tubes, including the ones used in this amplifier. They are based on tube SPICE models found at Norman Koren’s Vacuum Tube Audio Page and Duncan’s Amp Pages. Note: heater connections are not required for any of the triode models.

The two basic types of simulation included are frequency response (AC sweep) and time domain. The time domain analysis shows the shape of the output waveform and can be used to determine the amplifier’s harmonic distortion. They both run from the same schematic, but the input sources are different. For the frequency response simulation, the audio input is a VAC (AC voltage source). The time domain simulation requires a VSIN (sine wave generator) input. Before running a simulation, make sure that the correct AC source is connected to the amp’s input on the schematic.

The following instructions for using the simulation files are not a complete tutorial for OrCAD. The OrCAD HELP files and online manuals include tutorials for those who want to learn more about OrCAD.

Frequency Response (AC Sweep) Analysis

1. Run OrCAD Capture and open the project file “Morgan Jones.opj”.
2. In the Project Manager window, expand the “PSPICE Resources|Simulation Profiles” folder. Right click on “Schematic1-freq_resp” and select “Make Active.”
3. In the Project Manager window, expand the “Design Resources|.\morgan jone.dsn|SCHEMATIC1” folder and double click on “PAGE1”.
4. On the schematic, make sure that the input of the amp is connected to the V3 AC voltage source. If it is connected to V2, drag the connection to V3. By default, V3 is set to 0.5V. (Note: the tubes in the OrCAD schematic are labelled U1, U2 and U3. In the article schematics, they are referred to as V1, V2a and V2b.)
5. To add the triode library to the Capture: click the Place Part toolbar button (). The Place Part dialog appears. Click the Add Library button. Navigate to the triode.olb file and click Open. Make sure that the analog.olb and source.olb libraries are also listed in the dialog. Click the Cancel button to close the Place Part dialog.
6. From the menu, select PSpice|Edit Simulation Profile. The Simulation Settings dialog appears. The settings should be as follows:
   Analysis Type: AC Sweep/Noise
   AC Sweep Type: Logarithmic (Decade), Start Freq = 10, End Freq = 100K, Points/Decade = 100
7. To add the triode library to PSpice: Click the “Libraries” tab. Click the Browse button and navigate to the the triode.lib file. Click the Add To Design button. If the nom.lib file is not already listed in the dialog list, add it now. Then close the Simulation Settings dialog.
8. To display the input and output frequency responses on a single graph, voltage probes must be placed on the input and output points of the schematic. Click the Voltage/Level Marker () on the toolbar and place a marker at the junction of R9 and the grid of U1. Place another marker just above R_Load at the amp’s output.
9. To run the frequency response simulation, click the Run PSpice button on the toolbar (). When the simulation finishes, the PSpice graphing window appears. The input and output curves should be in different colors with a key at the bottom of the graph.
10. The PSpice simulation has computed the bias voltages and currents in the circuit. To see the bias voltages displayed on the schematic, press the Enable Bias Voltage Display toolbar button (). To see the bias currents displayed on the schematic, press the Enable Bias Current Display toolbar button ().

Time Domain (Transient) Analysis

1. On the Capture schematic, make sure that the input of the amp is connected to the V2 sinewave source (the default values are: VAMPL=0.5, Freq. = 1K, VOFF = 0). If it is connected to V3, drag the connection to V2.
2. In the Project Manager window, expand the “PSPICE Resources|Simulation Profiles” folder. Right click on “Schematic1-transient” and select “Make Active”
3. From the menu, select PSpice|Edit Simulation Profile. The Simulation Settings dialog appears. The settings should be as follows:
   Analysis Type: Time Domain(Transient)
   Transient Options: Run to time = 10ms, Start saving data after = 0ms, Max. step size = 0.001ms
4. To display the input and output waveforms on a single graph, voltage probes must be placed on the input and output points of the schematic. Click the Voltage/Level Marker () on the toolbar and place a marker at the junction of R9 and the grid of U1. Place another marker above R_Load at the amp’s output.
5. To run the time domain simulation, click the Run PSpice button on the toolbar (). When the simulation finishes, the PSpice graphing window appears. The input and output curves should be in different colors with a key at the bottom of the graph.
6. To determine the harmonic distortion at 1KHz (the sine wave frequency), harmonics in the output waveform must be separated out through a Fourier Transform. In the PSpice window, press the FFT toolbar button (). The PSpice graph changes to show the harmonics for the input and output waveforms. The input and output curves should be in different colors with a key at the bottom of the graph.
7. The fundamental frequency at 1KHz will have the largest spike. The other harmonics are too small to be seen at the default magnification. In the PSpice window, press the Zoom Area toolbar button () and drag a small rectangle in the lower left corner of the FFT graph. The graph now displays a magnified view of the selected area. Continue zooming in until the harmonic spikes at 2KHz, 3KHz, etc. are visible.
8. Harmonic spikes should exist for the output waveform only. The input is an ideal sine wave generator and has no distortion. To calculate total harmonic distortion, add up the spike values (voltages) at frequencies above 1KHz and divide by the voltage at 1KHz (the fundamental).

Additional Simulation Tips

• To change the value of any component on a schematic in the Capture program, double-click on the value and enter a new value at the prompt.
• The schematic included in the simulation files is for the optimized Morgan Jones amplifier with feedback. To simulate the non-feedback amplifiers without removing the feedback resistors, change the values of R8 to 100meg and R9 to 1 ohm (or 100 ohms as the grid-stop R_g).
• To measure the grid-cathode voltage of tubes (V_gk), use the Voltage Differential Marker (). Click the Voltage Differential Marker toolbar button and touch the probe to the tip of the grid pin and then cathode pin.
  

Note: simulations only approximate the performance of a circuit. The actual performance may vary considerably from the simulation as determined by a number of factors, including the accuracy of the component models, and layout and construction techniques.

c. 2000, 2002 Chu Moy.

Addendum

10/17/2000: Updated section on power supply.

10/19/2000: Added section on impedance converter.

10/24/2000: Charles King suggested rating the plate resistors at 1W.

4/9/2002: Major revision to article: expanded discussion of original Morgan Jones design and added sections on the optimized Morgan Jones amplifier schematics by Alex Cavelli. Also added section on simulating Morgan Jones circuit using OrCAD Lite.

6/3/2002: Added schematics and descriptions for power supplies by Bryan Ngiam and Rudy van Stratum (figures 12-14). They both built the optimized version of the MJ amp without feedback. Here are pictures of Ngiam’s and Stratum’s amps:

More information about Bryan Ngiam’s power supplies is available at Bryan Ngiam’s Home Page.

3/3/2003: The author updated the 6DJ8 model in the triode library used in the OrCAD simulations of the Morgan Jones amp. He also revised value of R4 from 150 Ohms to 120 Ohms for the optimized amps (figures 5 and 10). He writes:

I’m including a new triode library. I have modified the 6DJ8 model based on some other work that I was doing. The parameters are slightly different from the previous version and they are more accurate. You should use this library and replace it in the ZIP you’re making available on the website. Because of this better model, I am recommending that the plate resistor for the MJ be lowered from 150 ohms to 120 ohms. Closer to 1/Gm.

A Zero-Feedback SRPP-Input Headphone Amplifier.

by Simon Busbridge

I recently purchased the Sennheiser HD600 headphones second-hand and realised that their full potential could only be met with the highest quality amplifier. Hence this project! Before starting there were certain criteria I wanted to satisfy:

• no feedback
• low output impedance
• no possibility of nasty DC appearing at the output, which could damage the headphones
• safe operation

Output transformerless designs either use direct coupling or capacitor coupling. In the former case the possibility of DC appearing at the output is very real, either at switch on or when plugging the ‘phones in and out (when the common connection in the plug can short the right channel jack output). With capacitor coupling it is usual to use a delayed muting switch to prevent switch on and off “thumps”. Generally speaking, there are no turn on and off thumps on transformer-coupled amps because the current in the output valve tends to build up and decay slowly, a fade-in and fade-away type of effect. Muting circuits on other designs are fine, but a transformer does offer the ultimate in terms of protection (damage to the headphones and electric shock).

The Amplifier

Figure 1

So that really decided the form of the output – it had to a transformer. To get a low output impedance I needed to use quite a high step-down ratio (20:1); after all, the amplifier may be used with headphones of lower impedance that the 300 Ohms of the HD600. The output valve is the 12B4A which has a anode impedance of about 1000 Ohms, so the output impedance of the amplifier is about 2.5 Ohms, which is very low for a valve amplifier. You can use other valves here but watch that the anode impedance is not too high or the output impedance and bass-cut off frequency will both rise.

It is usual with power amplifiers to match the impedance of the loudspeaker to the impedance of the anode circuit using the equation turns ratio = square root (impedance ratio). Under these conditions, maximum power transfer takes place from the amplifier to the load. With headphones, the power involved is so small that it is not necessary to operate under such conditions. It is possible to make the turns ratio such that the output impedance of the amplifier is much smaller than the load. Power transfer is not optimum, but the output valve is tending to run under constant current conditions which has the advantage of lowering distortion.

The output power will depend on the impedance of the headphones; the amp gives about 10 V output before clipping. The gain of the output stage is 0.33, so quite a bit of amplification is needed before it. I am a fan of SRPP because it improves linearity, lowers output impedance and allows for a large voltage swing. The first stage uses a double triode directly coupled to the output valve, giving an overall gain of about 20 dB. The circuit is shown in figure 1. The 12B4A cathode resistor tends to run quite hot so use a 7 W component here.

There is no feedback from the output to the input. The gain can be increased slightly by decoupling the cathode of the SRPP input, or alternatively try a ni-cad battery bias. Although I use the 6072A for this stage, suitable alternatives are the 5965 and 12AT7. The 12AT7 is slightly less linear. The 6922 can also be used with a change to the heater wiring but the gain will be lower.

The 2.2 K Ohms resistor on the output of the audio transformers is there to give the output valve some load if the headphones are disconnected. Without it, the transformer can ring off load. The output impedance of the amp is so low that its value is quite uncritical.

The Power Supply

Figure 2

The power supply, shown in figure 3, is fairly conventional, with valve rectifier and choke input filter for quiet operation. An extra stage of filtering removes any residual ripple and helps to separate the channels. All sorts of different power supplies can be used, but it needs to be capable of 60 mA at 200V. The 200V HT voltage is not critical. DC can be used on the heaters, but it is not really necessary as the gain is fairly low. Solid state power supplies can be used. They just don’t sound as good!

Figure 3

With my Sennheiser HD600, the amp will clip if you try to go too loud – 100 dB or so. I actually like it that way, because it is very easy to damage your hearing with headphones without realising it. There is, therefore, an in-built tendency to back off the volume control slightly. To increase the drive capability, raise the HT supply by 100V or so. This can be done by either changing the transformer secondary winding, or by placing a smoothing capacitor directly after the rectifier (figure 3).

In the former case the power supply remains choke input, but watch the maximum rectifier voltage or the valve rating may be exceeded, because you lose quite a few volts with this method. In the latter case, it becomes capacitor input. With a choke input filter, the output voltage of the supply is 2/p V_p, where V_p is the peak output voltage. With a capacitor input filter, the output voltage is just V_p or about 300VDC. I cannot tell any difference in the sound quality – take your pick!

Construction

The complete system is built on two separate chassis to avoid magnetic field induced hum (I recommend that you ask for a flux-band and earth-screen to be fitted to your transformer). If it is kept on two chasses (amp on one, psu on the other), then component placement is quite uncritical. The chassis frames measure 9″ x 10″ x 4″. They are assembled from hardwood boards (deal/pine wood) with mitred corners. I used a mitre saw to make the cuts, which was not easy. A circular saw set at 45 degress is how I do it now. The joins are with the small brackets screwed through from the inside.

The top plate 1.6 mm thick sheet metal, drilled or punched (with Q-max punches) in the right places, and then pearl anodised at a local anodising factory. On the chassis for the power supply, the top plate has strengthening bar across the middle; otherwise the top tends to sag. The wooden parts can be spray painted black and then polished for good effect. The finish is excellent as the grain of the wood shows through.

The mains switch is a rotary type. The shaft is a bit of old 1/4-inch brass rod fixed on the switch with a standard spindle coupler and held in the front panel by a bush, again standard off the shelf. The method is cheap and works well. The switch itself is mounted on a small bracket, just a scrap of metal bent to 90 degrees with the necessary holes in it. For T1 in the amplifier, I used 20:1 line output transformers (part no. pa106 from SJS Electroacoustics). The power supply transformer and choke were custom-made by Sowter of Ipswich. I paid 165.59 UKP for the two.

The lamp is an integral diode, resistor and LED in a chrome mounting. You might wish to consider fitting a fuse at the input. Some people say that they can hear fuses! Here in the UK, I use a 2A fuse. In the USA, the current will be higher so I suggest a 5A fuse. A 5A fuse is also OK for the UK. All UK mains plugs are fused anyway (unlike the US) with a maximum of 13A.

The input phono jacks are located round the back, along with the umbilical between power supply unit and the amplifier. The Switchcraft headphone jack is on the front (left) next to the volume control (right). The plan was to have a gold-plated Switchcraft jack, which I saw at an AES show, but I could not get hold of one, so it is a standard tin-plated Switchcraft.

I have only given the power rating of resistors which dissipate any appreciable power. For all others, 0.5 W resistors will do. The capacitors came from all over the place. The 1600uF big one was second hand in a junk shop; the others from HAM fests.

The Result

The amplifier should work with headphone impedances as low as 32 ohms; I have tried lower impedance Sennheisers and they seem OK. I would be worried at less than 10 Ohms or so. The sound quality of the amplifier is excellent. The lack of feedback produces an open natural, detailed, sound without any of the harshness sometimes experienced with solid state amplifiers. I can now listen in bliss well into the night when other members of the household have gone to sleep! This is definitely one of my best projects.

All of my hi-fi equipment is built along the same lines as this headphone amp. I plan to get around to designing feet which go over the corners of the boxes from top to bottom to hide the mitre joins. The top will have a nice domed finish and the bottom adjustable spikes. But like all things, it is the time…

Dr. Busbridge is a lecturer in the School of Engineering at the University of Brighton (U.K.).

Addendum

6/7/99: Added paragraph discussing output transformer impedance when driving headphones.

3/25/01: Various revisions to article text. Added high-res images. The rectifier tube in the power supply (figure 2) was changed to a 5U4G. The GZ3 previously shown can also be used.

3/27/01: Changed value of first filter capacitor in figure 2 from 100uF to 1600uF.

c. 1999, 2000, 2001 Simon Busbridge.