A SRPP-Input Tube Amplifier For Headphones And Loudspeakers.

by Tony Frazer

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I built the prototype of this headphone amplifier in Oct’96 and remain impressed by it’s performance. Imaging is particularly good – I make much greater use of my headphones than ever before! Hum is low and unintrusive. I have used 60 ohm and 300 ohm headphones with this amp. The amp is designed also to drive an efficient pair of 8 ohm loudspeakers. If you do build it, please let me know how you get on – ideas for improvement are always welcome!

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Figure 1

Figure 1 shows the amplifier schematic for one channel. The circuit topology is a SRPP (Series Regulated Push-Pull) input stage which is AC-coupled to a Parallel-Triode Cathode Bias output stage. To check the circuit gain, I injected a 1kHz sine wave at 1V p/p into the grid of the lower V1. I measured 12V p/p at the grid of V2, and 110V p/p at the anode of V2. The output stage runs at 32mA, 234V, i.e. 7.5 Watts. Class A amplifiers are typically about 20% efficient, so it would be reasonable to expect about 1.5 Watts output.

The amp was designed to work with 8 ohm loudspeakers. Nearly all loudspeaker drive units I have encountered have a nominal impedance of 8 ohms – of course this varies with frequency by a few ohms. When driving headphones, the audio output transformer must be shunted with a 10-ohm resistor (Ro) to present the correct load to the transformer secondary. The shunt allows higher impedance headphones to be used. Remove the shunt (add a switch to take it out of the circuit) when driving loudspeakers.

A wide range of headphone impedances can be used with this amplifier because of its low impedance output. A pair of 60 ohm headphones will present 1/((1/10)+(1/60))= 8.57 ohm load with 60 ohm headphones, while the amp will see a 1/((1/10+(1/300)) = 9.68 ohm load with 300 ohm headphones. I know it isn’t efficient from a dedicated headphone amplifier perspective, where the output transformer would have a secondary matched to the headphone impedance. However, with the 10-ohm shunt, there is still plenty of power.

CONSTRUCTION NOTES

WARNING: Tube circuits involve potentially LETHAL HIGH VOLTAGES and should only be tackled by experienced persons with due regard for SAFETY. Don’t fiddle with the circuit while wearing your ‘phones!

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Figure 2

General construction notes for my prototype:

The unit is built on a two-section ‘U’-type Aluminium chassis WxDxH:12x6x2″ with holes in the base section and around valveholders for ventilation.

  1. Chassis mounting B9A valveholders are used throughout.
  2. Point-to-point wiring is used – including a couple of tagstrips.
  3. Mains and output transformers are situated at opposite ends of the chassis, mounted on butyl spacers to suppress vibration which may be picked up by the input valves, with the mains transformer at 90 degrees to minimise magnetic coupling.
Chassis plate layout:
+-------------------------------+  Key:
|         [h] [h]  [V2] [o/p ]  |  tr = Mains Transformer
|[   ]    [c] [c]       [    ]  |  R = rectifier tube
|[tr ][R]              [V1][V1] |  h = heater supply capacitor
|[   ]      [C]         [    ]  |  c = 100uF HT capacitor
|                  [V2] [o/p ]  |  C = 330uF HT capacitor
+-------------------------------+  V1 = 7025; V2 = 12BH7
                                   o/p = SE output transformer

AMPLIFIER CIRCUIT NOTES:

 

  • V1 is a Sovtek 7025/12AX7WA (i.e. a high spec ECC83). I tried 6072 (12AY7) and ECC81 types (with minor bias adjustments). ECC81 sounds a bit sterile, ECC83 has warmth! Raytheon 6072 was also OK, but past experience suggests they may go microphonic.
  • V2 is 12BH7 – I use a couple of Brimars, which, oddly enough appear to have slightly different internal structures! Although they both sound the same, this is something to watch out for if symmetry is important to you!
  • VOL and BALANCE controls are dual gang pots.
  • BALANCE slider is wired to input side for one channel and VOL side for the other. In use, VOL up to half-scale with a CD player is about the limit.
  • All resistors are 0.6W Metal Film unless otherwise stated.
  • 100nF coupling capacitor is polypropylene.
  • Screened leads link Input socket-BAL-VOL-V1 and ground at the Input socket.

Output tranformer is a home-made 2k7-8R using dual isolated bobbin for safety. The ‘1’ on the schematic denotes start (inside) of the windings. Primary: 800t of 32g; Secondary: 45t of 22g. I used a 20VA transformer kit to source the laminations and bobbin. Everything you need to make all the transformers can be obtained from Maplin Electronics by mail order. From the catalogue:

Stock Number / Description
YJ61R / 20VA Transformer Kit
YJ62S / 50VA Transformer Kit

Maplin Electronics also do copper wire on 50g and 250g reels in the gauges required (I would suggest the 250g reels). Heavy items such as transformers have an additional carriage charge.

Maplin Electronics
PO Box 777
Rayleigh
Essex SS6 8LU
United Kingdom

Laminations 2 5/8″ x 2 1/4″; 7/8″ stack. I use a variable-speed hand drill and a coach-bolt/wooden block which the bobbin fits over. On slowest speed, I count revolutions for a minute, then calculate winding time from that. Secondary is wound manually. Don’t interleave ‘E’ and ‘I’ laminatons as you would with a power transformer. No extra gap spacer is required as the ampere turns are well within limits for DC saturation. I use plenty of 1-hour epoxy resin to hold it all together. (That was not intended to be a lesson in winding transformers!)

  • Heater wiring not shown – see Power Supply schematics (figures 3 and 4) for heater supply. All heaters connected in parallel: connect pin 4 to pin 5, centre is pin 9.

 

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Figure 3

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Figure 4

NOTES ON THE POWER SUPPLY:

There are two versions of the power supply: the original prototype’s power supply (figure 3) and an all-silicon version (figure 4). The original uses a half-wave thermionic rectifier (EZ81). The all-silicon power supply uses a different transformer HT secondary, and the rectifier heater winding is not required. I have shown a capacitor-choke-capacitor filter which should prove quieter than the original resistor-capacitor chain filter.

Basically, the PSU provides +234V DC for the amp and 6.3V DC for the heaters, but because I am using an SRPP stage where the cathode of the upper half of V1 is at quite a high voltage with respect to gnd, I use the voltage-divider network to ‘pull up’ the heater voltage (with respect to gnd) to reduce the potential difference between cathodes and heaters.

Some tubes have a specified design limit – I got into the habit of doing this with SRPP stages using the 6072 tube for which a maximum of 90 volts K/h is specified, however the ECC83/12AX7 has a maximum ‘peak’ K/h of 200 volts specified – it is arguably unnecesary for the ECC83, but for the cost of a couple of resistors and a cap. it seems a worthwhile precaution to prolong tube life, or when experimenting with different tubes.

A green L.E.D. and 1K5 resistor (not shown) are connected across the smoothed heater supply as a pilot light.

Addendum

11/9/98: Corrected resistor value from 47 to 47K ohms in figures 4 and 5.

c. 1997 Tony Frazer.

Blue Hawaii Hybrid Electrostatic Amplifier for Stax Omega II Headphones.

by Kevin Gilmore
(Project Editor: Chris Young)

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The Blue Hawaii amp is my latest design in my search for the perfect amp to pair with my Stax Omega II headphones. The genesis for this hybrid electrostatic headphone amplifier occurred when I was in Hawaii on vacation, at a fancy hotel on Maui. Sitting at the bar on the beach, drinking “Blue Hawaiis,” I drew the schematic for the amp on a placemat. The design is my conception of the mysterious and rare Stax T2 amp, which I have never been able to find at anything resembling a rational price.

I searched out any information I could find on the T2 in an attempt to create my own version. I was able to determine that it used EL34s as output tubes in a grounded grid configuration, which is the lowest distortion tube output circuit known. It also used 6DJ8s as input tubes with some solid state in the second and third stages. My design uses the first and second stages from my solid state electrostatic amplifier coupled with a third FET stage and then the final grounded grid stage.

My design ended up with a fairly large amplifier pulling significant amounts of power which results in a very smooth and extended frequency range from DC to over 200khz (-3dB at 400khz). Of all my electrostatic amps, this one has the largest output voltage swing. This is not an amplifier for the timid, nor is it a good idea to build this as your first project, though some, however, have actually done so.

The Circuit

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Figure 1
(Click here to see single-image schematic of amplifier.)

Figure 1 is the amplifier schematic. The entire amplifier has a differential topology from input to output to get a balanced input and for lower noise, less ground loop problems. The first stage is a differential amplifier with feedback directly from the output stage. It works equally well with both balanced and unbalanced audio input sources. The step attenuators from Goldpoint make good volume controls for this stage. The JFET device (Q1) is a dual JFET all on one wafer. It is known for extremely low noise and excellent matching, and is used in a number of expensive designs, such as the Nelson Pass amplifiers. Q17 is a current source that sinks 3mA.

Because the amp is totally DC coupled from input to output, drift in the input stage is a bad idea. Since the first two stages run in current mode, the JFET input is more linear than a pair of bipolar transistors. Dual transistors all on one wafer suitable for audio use are hard to find these days. The FETs steer current away from the current sources Q2 and Q3. Together Q2 and Q3 each supply 2mA or a total of 4mA. The Q17 current source takes away 3mA leaving 0.5mA in each of Q4 and Q5, but some of the sink current is coming from the output feedback, so each FET is actually using somewhere between 0.5mA and 1mA.

The approximate voltage gain of this stage is 5; this stage really runs in current mode. The unit was designed to work equally well in both balanced and unbalanced mode. For single-ended signals, ground either the + or – input and apply signal to the other. The much higher impedance of the JFET works better when one side is grounded for unbalanced inputs.

The second stage starts with a constant current source (Q2 and Q3). The current source feeds a common base amplifier (Q4 and Q5). The common base amplifier feeds a modified Vbe multiplier. I believe a famous designer is now calling this circuit a current tunnel. It’s the most linear way of translating the voltage down to the bottom rail. The voltage gain of this section is about 4. The basic idea of the first two stages is to supply the third stage with a very fast low impedance drive signal that is referenced to the bottom rail.

The currents flowing into the common base amplifier (Q4 and Q5) are the difference between what Q2 and Q3 are supplying and what the FET is taking away. The rest of the current goes down the tunnel to the vbe multipliers (Q6 and Q7) that convert the current back to voltage. The current sources in the second stage supply 2 mA each. With no signal, the FETs take 1 mA, leaving 1 mA going through the common base amplifier into the bottom transistors, which are wired as Vbe multipliers (like a zener diode in series with a resistor, except a lot less noisy). This generates the 13 volts (referenced to – rail) necessary to properly bias the third stage.

The third stage is another differential amplifier (Q13 and Q14) being driven via another constant current source (Q10 and Q16). The voltage gain is about 200. Q11 is the power supply for this stage and makes a 100 volt power supply with -400V as the reference. The power supply voltage for this stage is kept down to 100 volts to reduce the Miller effect and keep the frequency response up. The higher output impedance of this stage is lowered by the use of 2SJ79 transistors, which are used as zero voltage gain emitter followers. The use of FETs in this stage coupled with the current source further reduces the distortion and provides for a solid low impedance drive signal for the output stage.

The 4th and final stage is a tube in grounded grid configuration (V1/Q8 and V2/Q15), similar to the common base amplifier in the 3rd section of my solid-state current-domain electrostatic amp. Q9 and Q12 are high compliance current sources and supply 25mA of bias current. Think of them as linear pull-up resistors for current (in fact, one builder has replaced the current sources with large resistors). The use of a current source here instead of load resistors acts to further linearize the output stage and reduce output distortion. V1 and V2 are the equivalent of common base amplifiers and do the entire rail-to-rail output voltage swings.

With feedback, the overall voltage gain of the amp is exactly 1000. The frequency response is kept high due to the low impedance cathode drive. The EL34s are biased at 10 watts and have an 800V voltage swing (by comparison, the output tubes of my original DC-coupled electrostatic amp are biased at 2 watts with a 600V swing), resulting in a frequency response well in excess of 100kHz into a 150pF load. (+0/-0.1dB).

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Figure 2

A regulated power supply design is shown above. The ±15V supply is made with the standard 7815/7915 regulators. The high voltage supply is a pair of 400 VDC supplies, glued together at the output (P-channel MOSFETs are a lot more money than the equivalent N-channel MOSFET). In each section, beginning with a 460V raw supply, a PNP transistor (2SA1968) is used as a current source to feed the 400V zener reference. Then a N-channel FET is used as a high impedance, input voltage follower and outputs 400VDC. By the way, the same exact supply with a 350V zener reference string instead and a slightly smaller transformer (without filament windings) is what I use now for the solid state current domain headphone amp.

The bias supply is a voltage doubler with an adjustable reference. It has a range of about 350VDC to 650VDC. For low bias Stax headphones, put a 10M resistor to ground at the end of the 4.7M. to make the output voltage .66 times the voltage before the 4.7M, which puts it in the range for low bias.

Construction

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(Click here to download pc board patterns in pdf format. 1)
(Click here to download pc board patterns in pdf format. 2)
(Click here to download pc board patterns in pdf format. 3)

Caution: This project involves working with high voltages, so be extremely careful! Keep one hand behind your back at all times. 800VDC across both arms might possibly stop your heart.

This amp was assembled on three printed circuit boards (two for each channel of the amp and one for the power supply) and housed in separate enclosures. A complete set of pc board patterns (pdf format) can be found above. They could be sent to just about any circuit board manufacturer to have boards made. The top of the board is almost all groundplane. All the parts, including the tubes, are mounted on these boards – the tubes are installed in pc-mounted ceramic tube sockets from Parts Express. The tubes must be exposed through the chassis. They dissipate 20W each (actually 10 to 12 watts of plate dissipation plus another 6.3V * 1.6A = 10 Watts of filament power).

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It was so much easier to do a pc board for this amp, but if I were to make a prototype, I would again use the 0.1mm perf board; the layout look much like the circuit board. (Note: For a layout in a single chassis, see the interior view of Headamp.com’s Blue Hawaii amp below.) 99% of the wiring would be on the bottom, and it would be, therefore, rather flat. Mounting the tubes would be trouble though, and would cause mechanical problems. The tubes are fairly heavy and get stinking hot. Each chassis measures 12″ x 10″ x 3.5″. (Note: Headamp.com is selling the Blue Hawaii design in a single chassis measuring about 16.5″ W x 16.0″ D x 3.5″ H and may sell the Blue Hawaii pc boards. Please contact Headamp for more information.)

I have Mullard EL34 tubes, but keep them put away due to what I could sell them for if I wanted. I actually used the National Union tubes from Richardson Electronics which cost $11.50 US each.

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All of the parts except the 2SA1968 have lots and lots of sources such as Digi-Key, MCM Electronics and Mouser Electornics. Only B&D; Enterprises has the 2SA1968 in the United States. In Japan and Canada, they can be ordered from Sanyo direct – the minimum order is 100 at a time, but then they are $1.25 each or so.

Q9 and Q12 are each made of six 2SA1968 transistors in parallel with one 2SA1968 as the driver. Matching the transistors is not required – unless one of the 2SA1968s is way off compared to the rest in which case it might get way too hot.

All resistors are 0.25W except where labeled. It is important to have all the pnp current source transistors correctly mounted to a large heatsink with silicon impregnated washers. If any one pnp transistor gets too hot it can short out the whole current source.

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Standard tab heat sinks will do for the 2SK216 and 2SJ79 transistors, but the 2SA1968 and 2SC3675 transistors must be mounted a big heatsink (one for each channel), capable of dissipating 20 Watts of heat. I obviously fabricated them, but otherwise they can be obtained from Conrad Heatsinks cut to length. The IRFBC30 MOSFETs in the power supply must be heatsinked too: Mouser part number 532-529902b25.

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The Stax SRC-5 headphone jack came from AudioCubes.com. Since the price has gone up to $19 each (I paid $10), it may be more cost effective to use the Allied jacks (see the current domain amp project article). Allied has a $25 minimum order, the cost of three pieces. Then they must be filed down on a lathe. Actually I am buying the male connectors from Allied, because no one else sells them. The male connectors are much easier to convert to standard Stax plugs. The power supply-to-amplifier connectors are the Amphenol military 12-pin connectors. The 4 connectors (two male and two female) were $130.

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The custom Victoria Magnetics power transformer has these specs: 2 x 330VAC/150mA, 36VCT/100mA, 2 x 6.3V/5A (filament supplies). Everything with Victoria Magnetics is custom. I paid $110 for the transformer with shipping. They know about the Blue Hawaii design and will supply the correct transformer on request. For safety, I recommend a 2A/110Vac fuse located in the input line to the power supply.

Setup and Results

Test voltages (with the amp at idle) are shown in red on the schematic and are with respect to ground. To set up the amp, adjust the two pots in each channel of the amp. P1 controls the differential output voltage. Put a voltmeter between the 2 stators of one channel of the headphone and set this pot for zero. P2 controls the voltage with respect to ground. Put a voltmeter between any stator and ground and set for zero. Then repeat both adjustments a few times. The plates of both tubes should measure 0 volts with respect to ground when the pots correctly adjusted. Once the pots are adjusted, that’s it – there’s no change from headphone to headphone.

Setting the bias voltage depends on the headphones. For Stax headphones that can accept a high bias voltage, adjust the pot for 560V. I do not think that the Omega II headphones can be damaged by this amp unless the bias is set way too high. If the bias is set right, the outputs are close to 0V at idle, and all the LEDs are lit, then the amp pretty much has to be working correctly. Now if one or more of the outputs is stuck at +400V or -400V, then something is seriously wrong and needs to be fixed. An oscilloscope really helps.

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Adjust the pot to 580 volts for Sennheiser HE-90 and HE-70 headphones or leave it at 560V. For Koss headphones, adjust the bias for 600V. To use the Sennheiser HE60 headphones with this amp, I made the adapter shown above. Those are RS-232 female connector pins that fit the HE60 pins perfectly. By hand I cut a circuit board with lands exactly 3.5mm apart put the pins on the HE60 connector, lay them down on the circuit board and solder. Then attach wires and a standard stax plug.

The amp can output 1500 V p-p measured stator to stator. At 800Vp-p, THD is less than 0.004% from 20Hz to 20kHz. The actual frequency response is 0 to 100kHz (-3dB at 150kHz) into an Omega II load. Compared to the sound of my previous tube amplifier, the bass is no longer tubby; it’s very sharp and tight. The high end is no longer rolled off, so female voices sound much more real. If the bias supply is reduced to 280V, the amplifier will drive all electrostatic headphones. I tried it last night on a pair of SRX’s. I never ever heard them sound so good.

Previously with a standard dummy head, I measured the SPL in Omega 2 headphones driven by this amplifier. With a drive signal of 800Vp-p per side, the resulting spl is 106dB. THAT’S LOUD! The amp can put out 1500 volts peak-to-peak, and thats louder! I just ordered a pair of Stax SR-001 MkIIs, which can reach up to 120dB. My ears distort before the amplifer/headphones do. It is quite loud at clipping, but the clipping is a hard clip with no oscillation or ringing. To use the amplifier with electret headphones, delete the bias voltage. And probably keep the output swing under 200V. Electrets phones when driven with this amplifier can probably get very very loud.

Several of my previous electrostatic designs are available in the Headwize Projects section. Comparing the Blue Hawaii to my all solid-state current domain amplifier, they really are more the same than they are different, but in general, the differences are the differences between tubes and solid state, such as a much smoother high frequency response, which in the case of the Blue Hawaii goes well beyond 500kHz. Additionally, the four times power consumption of the Blue Hawaii means a much stiffer and tighter bass response. Even though both are flat to zero and test similar, the BH bass is much more apparent and tighter.

c. 2004 Kevin Gilmore.

A Simple Tube/Opamp Hybrid Amplifier.

by Alex Cavalli, Mark Lovell and Bill Pasculle

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Introduction

After seeing many of the excellent and eye-catching tube-solid state amplifiers of others, we’d like to present a slightly different topology of hybrid amplifier design, using the same two basic components of a tube and an opamp. This amp is simple and has less than $50.00USD worth of parts and no lethally high voltages. It makes an ideal tube headphone amp for those who like the sound of tubes but worry about the safety of high voltage equipment.

Bill was looking for an amp that sounded good and also was a good companion for his Rio Carbon portable player. He also suspected that others might like something similar as well. With high bit-rate MP3 or WMA files, the Carbon can produce excellent sound, but as with many other portable players, benefits by the addition of a decent headphone amp. He also wanted to use the amp in his cubicle at work – requiring good sound without taking up too much space.

The design of the amp started when Bill suggested that he wanted to build one of the YAHA (Yet Another Hybrid Amplifier) amps , but using a different design. Alex, being mostly interested in tube amps, was not initially interested but as Bill and Mark began to suggest requirements for the amp and some options for the input tube, it was obvious that there was room to improve the typical hybrid topology. After more thought, Mark provided a list of some requirements for the design that he felt were necessary to meet in order to improve on the hybrid topology.

Alex went to work taking the design discussions and turning them into a draft design. Our first name for the amp (as a joke) among the team was Stoopid Opamp Headphone Amp (SOHA). The name, as so often happens with skunk-works project names, eventually stuck and finally we are just calling this amplifier the Stoopid or the SOHA. Another name for the amp, with the same acronym, might be the Simple Opamp Hybrid Amplifier.

The original prototypes assured us that it is possible to make a surprisingly good performing amp utilizing a tube at relatively low voltage and while still keeping the build cheap, easy, and reasonably electric shock-free. After altering some of the power supply and circuit values and testing the prototype, we ended up with something that was stable and fun to listen to for extended periods. It applies some compression but that’s part of its charm, and it would be a rather sterile sounding amp without the 12AU7/ECC82 altering the sound the way it does with its 40V plate voltage.

Amplifier Circuit

Most of the hybrid amps that have appeared in HeadWize threads and elsewhere (such as the Millet hybrid) have used the same B+ for both the tube and the opamp. Some of these amps are designed to be portable enough to run from a battery, but most are really constrained to a low voltage DC supply of some kind plugged into the line and so are not truly portable.

In addition, it is generally true that tubes that are not designed for low voltage use will not perform well at 12-24V (which is why the Millet uses special low voltage tubes), so we decided to try to provide the tube with higher B+ to get better performance, while still keeping the voltages fairly low. This meant that the amp could be small and portable although requiring AC power. Like the other hybrid amps, the SOHA is designed to give the sound of tubes while avoiding the high voltage risk that some builders don’t like. Still, providing a higher B+ permits us to get excellent sound from a more commonly available tube like the 12AU7/ECC82, which is in good supply from NOS and current production sources and which gives a wide variety of choices for tube rolling. Having this wide selection also makes it easier for the amp to be constructed in any part of the world.

The way in which most other hybrids use a common B+ for both tube and opamp has two detrimental effects on a hybrid amp:

  1. It puts the opamp in a single ended configuration where it needs an output cap to block half the B+
  2. It forces the B+ on the tube to be low so as not to exceed the maximum opamp rail voltages.

The first design decision was to decouple the B+ for the tube and opamp. Doing this makes it possible to use a standard bipolar supply for the opamp, eliminating the large output cap (as in the Chu Moy’s pocket amp for example) and requiring only a small coupling cap between stages (see power supply discussion below).

The tube is loaded with a constant current source (CCS) for two reasons:

  1. The resulting non-linear performance associated with low voltage operation is partially offset by the high dynamic impedance of the CCS
  2. A CCS has a much better PSRR than a simple resistor making it possible to have more ripple in the B+ and, therefore, simplifying the B+ PS.

The original amp is designed to run with FET input opamps. See note 1 (the BJT opamp section) for a version using BJT input opamps.

After extensive prototyping by Mark and Bill, and after posting the first design to the HeadWize forums and receiving feedback from several builders, we modified the original design. The most notable change was to the heater circuit. Originally the heater voltage was supplied directly from the AC secondary with voltage dropping resistors. This approach was implemented initially to maintain simplicity. However, because there is so much variation in transformer regulation, line voltage, and heater characteristics, the simple resistors were replaced with a regulator circuit. A side-benefit is the elimination of power wasted as heat since the dropping resistors reached 105°C – 115°C under normal operating conditions.

The Amplifier Circuit

The basic amplifier circuit uses a very small number of components, as shown:

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R1 100k Log Pot C1 1000u 10V Electrolytic
R2 300R 1/8W C2 100n 100V
R3 560R 1/8W D1, D2 1N4148 or similar
R4 2k Trimpot U1 OPA2134 or similar
R5 1M 1/8W V1 12AU7/ECC82 or equivalent
R6 150R 1/8W

Figure 1 – Basic SOHA Amplifier

The topology of the amp is a simple grounded cathode gain stage coupled through a capacitor to an opamp wired in unity gain mode. The standard SOHA uses LND150 depletion mode MOSFETs for the CCS for reasons discussed below. The amplifier circuit is, thus, very simple. Trim pots are provided as part of the cathode bias resistors to adjust for variations in tubes to set the plate voltages to ~40V. Each CCS is set to regulate at approximately 1mA.

An advantage of this design over many of the other hybrid designs is that there is no large coupling electrolytic at the output. The required inter-stage coupling capacitor is small making it possible to use good quality film/audio capacitors here at nominal additional cost.

With a 12AU7/ECC82 the input stage has a gain of about 12. This is sufficient for almost any source driving almost any headphones which is why the opamp is simply operating as a unity gain current buffer. However, with such high gain it is possible to exceed the input voltage tolerances for the opamp with just 1Vp at the input. The data sheet for the OPA2134 (and many similar opamps) indicates that the maximum input voltage is (V-) – 0.7V to (V+) + 0.7V. This means that the input swing must not exceed the supply voltage by more than one diode drop. The diodes ensure that this does not happen.

If the diodes conduct, the excess current passes into or out of the bipolar power supply. What happens thereafter depends on the ability of the tube to source/sink current and the ability of the PS to sink/source it. In this case, the tube will source/sink in the range of hundreds of micro amps which will find their way to the output caps of the bipolar supply which are in turn supplying current to the opamp V+ and V-. The output of the regulators will fluctuate some, but at this point the amp would not be operating properly anyway.

The standard CCS for the basic SOHA uses a single LND150 MOSFET in this configuration:

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R7 1k 1/8W R8 360R 1/8W

Figure 2 – Standard LND150 MOSFET CCS

For a discussion of why this was selected as the standard CCS, see note 2 (the CCS comparison section) at the end of the article. One limitation on the standard SOHA CCS is the uneven availability of the LND150 MOSFETs globally. To ensure that this amp can be built almost anywhere, we have created two variations that use other devices for the CCSs. The first uses J113 JFETs and the second uses 1N5297 current regulator (CR) diodes. Here are the schematics for both variations:

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R8

1k5 1/8W

Figure 3 – Alternate CCSs using JFETs and CRDs

Care should be exercised when building the SOHA with the J113 JFETs. Their maximum Vdss is 35V. Under normal operating conditions they will see only about 15-20V, but if the plate voltage on the tube is too low it is possible to exceed this maximum and destroy them. To protect the JFETS, the minimum plate voltage should never be set below 20V (see below the warning about adjusting the trimpots). The 1N5297 CRD has a 100V maximum and should withstand all of the normal voltages in this amp. The J505, noted in parenthesis, will also work but has only a 50V maximum. This makes the J505 a little more robust in this circuit than the J113, but less desirable than the 1N5297.

Power Supply Circuit

The key to this amp is the power supply. Initially the amp used an easy-to-acquire 30VCT/200mA transformer. As noted above, during the development and testing process, including builds by several HeadWizers, we changed the heater supply from AC to regulated DC. To accommodate the 150mA DC drawn by the heater it is necessary to increase the current spec on the secondary to 400mA. This will also give some headroom for the amp itself. Eventually, we chose the Amveco TE70053 toroid to replace the original split bobbin transformer. Another benefit to using the toroid is less EM radiation in the box and, since the SOHA also designed to be small, this reduces or eliminates problems with PS buzz. Other transformer possibilities are in the Power Supply section below. You can use a higher current rating transformer without difficulty, but if you increase the voltage be careful about not exceeding the maximum input voltage for the regulators. The bipolar opamp supply is a conventional regulated supply using 78L12/79L12 inexpensive regulators. They have a maximum input voltage of 40V.

The trick to the power supply is the use of a 1.5x full-wave voltage multiplier to generate the B+ for the tube. To make the voltage multiplier, the entire secondary of the transformer is rectified through a pair of coupling capacitors and bootstrapped on top of the V+ of the bipolar supply. With a typical transformer with 25% regulation and with no load on the B+ for the tube, this generates over 80V (this is marginally dangerous and will give you a pretty good sting so be careful).

The power supply, including the heater circuit is shown below:

cavalli2_fig4.png

R9 2k2 1/8W BR1, BR2 100V 1A Bridge Rectifier
R10 1k3 1/8W VR1 12V Fixed Regulator 78L12
R11 11k 1/8W VR2 -12V Fixed Regulator 79L12
C3-C6 100u 100V VR3 Adj. Negative Regulator LM337
C7, C8, C11 47u 16V D3, D4 1N4002
C9, C10, C12 470u 35V T1 30VCT 400mA Transformer

Figure 4 – the SOHA Power Supply

When the B+ is loaded with the tube, with each triode drawing ~1mA, the voltage is pulled down to between +55-65V. This means that there is plenty of headroom in the B+ to run the tube at+40V while still leaving space for driving 7-10V into the opamp. And this seems to give very good performance. As noted above, using a CCS for the plate load relieves ripple requirements on the B+ so a much simplified and less expensive filter section becomes possible. For the components as drawn the B+ ripple is about 1mV. The capacitor values are kept low and, hence, the capacitors are small and inexpensive. Again, for CCS PSRR comparisons see below.

The heater supply uses a full-wave rectifier into a negative 12.6V regulated supply. Pay careful attention to the orientation of the rectifying diodes. The heater supply is attached to the negative half of the bipolar supply. This was done because the heater current will pull down the input to the filter section of whichever half of the bipolar supply to which it is attached. Since we are using the positive supply to bootstrap the B+ for the tube we don’t want the heater supply to pull this voltage down. Therefore, it is derived from the negative supply because if the negative input to filter drops by a volt or two the regulator will not be affected. Pay careful attention to the orientation of the rectifier diodes and capacitors in the heater circuit since it is a negative supply. A power indicator LED can be attached to the heater supply taking care to note the polarity. The negative regulator should be heatsunk to dissipate about 2W.

Construction

The full schematic for both channels and the PS is shown below with the complete parts list less some miscellaneous components such as enclosure, power switch, etc.

cavalli2_fig5.png
Click here to see full-size schematic.

R1 100k Log Pot C3-C6 100u 100V
R2, R12 300R 1/8W C7, C8, C11 47u 16V
R3, R13 560R 1/8W C9, C10, C12 470u 35V
R4, R14 2k Trimpot D1, D2, D5, D6 1N4148 or similar
R5, R15 1M 1/8W D3, D4 1N4002
R6, R16 150R 1/8W U1, U2 OPA2134 or similar, dual or single
R9 2k2 1/8W V1 12AU7/ECC82 or equivalent
R10 1k3 1/8W BR1, BR2 100V 1A Bridge Rectifier
R11 11k 1/8W VR1 12V Fixed Regulator 78L12
R7, R17 1k 1/8W VR2 -12V Fixed Regulator 79L12
R8, R18 360R 1/8W VR3 Adj. Negative Regulator LM337
C1, C13 1000u 10V T1 30VCT 400mA Transformer
C2, C14 100n 100V

Figure 5 – Full SOHA amplifier, both channels and PS

For the J113 version eliminate R7, R17 and change R8, R18 to 1k5 1/8W. For the 1N5297 version simply replace the entire CCS with the single diode.

The amp has been built several ways by different HeadWizers [click here to see forum member Neurotica’s (Jim Eshleman) SOHA build narrative]. Mark and Bill initially built the prototypes using point to point wiring on perfboards and several others did so as well. Bill eventually also made a homemade PCB while Alex designed a PCB using the commercial package ExpressPCB (see below). Most builds to date have been like the prototypes with the PSU and amplifier circuits on the same board. Pictured below is a pictorial drawing showing how the SOHA can be wired point-to-point on a 4 x 6-inch perfboard.

cavalli2_fig6.png
Figure 6a – Bill’s SOHA constructed by point to point wiring on a perf board
Click here to see full-size layout.

Part of our purpose with the design and component specs is to keep everything as small and cheap as possible. The parts list shows the parts from the usual American suppliers. Mark was able to source similar parts from Farnell and RS in the UK.

Part # Mouser Catalog Number Description Qty. Price Total
R9 270-2.2K-RC Xicon 2.2k 1/8W

10

0.11

1.10

R3, R13 270-560 Xicon 560R 1/8W

10

0.11

1.10

R11 270-11K Xicon 11k 1/8W

10

0.11

1.10

R10 270-1.3K-RC Xicon 1.3k 1/8W

10

0.11

1.10

R5, R15 270-1.0M-RC (regular CCS) Xicon 1.0Meg 1/8W

10

0.11

1.10

270-100K-RC (mu follower) Xicon 100k 1/8W

10

0.11

1.10

R2, R12 270-300 Xicon 300R/1/8W

10

0.11

1.10

R6, R16 270-150-RC Xicon 150R 1/8W

10

0.11

1.10

R4, R14 652-3306K-1-202 Bourns 6mm 2K pot

2

0.56

1.12

C7, C8, C11 140-HTRL16V47 Xixcon 47uF/16V

3

0.07

0.21

C3, C4, C5, C6 140-HTRL100V100 Xicon 100uF/100V

4

0.42

1.68

C9, C10, C12 140-HTRL35V470 Xicon 470uF/35V

3

0.3

0.90

C1, C13 140-HTRL16V1000-TB Xicon 1000uF/16V

2

0.25

0.50

C2, C14 1429-1104 Xicon 0.1uF (100nF)

2

0.44

0.88

CCS Options
512-J113 J113 JFET

4

0.24

0.96

R8 270-1.5K Xicon 1.5k 1/8W

10

0.11

1.10

OR
689-LND150N3-G LND150 MOSFET

2

0.55

1.10

R8, R18 270-360 Xixon 360R 1/8W

10

0.11

1.10

R7, R17 270-1K-RC Xixon 1k 1/8W

10

0.11

1.10

  OR
610-1N5297 1N5297 CC Diode

2

4.29

8.58

BR1, BR2 821-DB102G 1A 100V Bridge

2

0.33

0.66

VR1 512-LM78L12ACZ LM78L12 Regulator

1

0.27

0.27

VR2 512-MC79L12ACP LM79L12 Regulator

1

0.34

0.34

VR3 512-LM337T LM337 Regulator

1

0.5

0.50

567-273-AB Wakefield Heatsink

1

0.38

0.38

Power LED 351-3310 Xicon Blue Led 3mm

1

1.5

1.50

271-560-RC 560R 1/4W LED Resistor

10

0.09

0.90

D1, D2, D5, D6 78-1N4148 1N4148 Diodes

10

0.03

0.30

D3, D4 512-1N4002 1N4002 Diodes

5

0.10

0.50

R1 313-1240-100K Taiwan Alpha 12mm pot, 100k

1

2.84

2.84

575-393308 IC Socket

1

0.36

0.36

J3 161-3502 3.5mm Headphone Jack

1

0.92

0.92

J1 161-1052 RCA Jack Black

1

0.82

0.82

J2 161-1053 RCA Jack Red

1

0.82

0.82

Total

38.04

OR
DigiKey
U1, U2 OPA2134PA-ND OPA2134PA

1

2.63

2.63

T1 TE70053-ND Amveco 30V CT 500mA

1

12

16.22

Total

18.87

Optional Sources
Newark
U1, U2 75C4624 OPA2134-PA

1

2.37

2.37

18C6948 J113 JFET

2

0.2

0.40

VR1 34C1091 LM78L12ACZ Positive regulator

1

0.28

0.28

VR3 34C1076 LM337T Regulator

1

0.67

0.67

R4, R14 46F1092 Bourns 6mm 2K pot

2

0.27

0.54

Antique Electronic Supply
V1 T-12AU7-JJ JJ 12AU7 Tube

1

8.95

8.95

V1 P-ST9-511 Tube Socket

1

1.95

1.95

Miscellaneous
Knob
Power Switch
Wire
Fuse holder and Fuse (0.25A)

The Amveco toroidal transformer (30VCT/500mA) is available from Digikey. Remember with 15-0-15 VAC (nominal) secondaries and the poor regulation of these inexpensive transformers, you will see over 21V with no load at the inputs to the bipolar power supplies and ~85V with no load for the B+. Because of the poor regulation make sure to use capacitors with voltage ratings that meet these off-load conditions. Note that the PS parts table shows 100V capacitors for the B+ section. If you use a higher voltage transformer make sure that the input to the regulators does not exceed their maximums (typically about 37V).

Some other possible split bobbin transformers are: Dagnall D3019 (0-240 pri), D3023 (0-115,0-115 pri). Both are 12VA. Other possible toroids are: MULTICOMP MCTA015/15 (0-115,0-115 pri), MULTICOMP MCFE015/15 (0-230V pri), or MULTICOMP MDCG015/15 (0-230V pri).

This design is optimized for 12AU7/ECC82 and its exact equivalents (5963, 6189, and 6680) rather than a close equivalent (or other types of tubes such as 6922).

All three flavors of CCS provide a degree of PSRR and some immunity from power fluctuations. They differ in availability worldwide and in maximum voltage ratings. The best overall CCS uses the LND150 MOSFET which is not available everywhere. The J113 FET is widely available but its maximum DC voltage is only 35 volts. Normally the FET wouldn’t see more than 15-20V unless the plate voltage gets too low. The 1N5297 CRD has a maximum voltage of 100 VDC but is not as widely available and is also expensive. Nevertheless, working amps have been built using all three types of CCS. Trim pots located at the cathodes are used to adjust the plate voltage. In order to prevent burning out the CCS FETs the cathode trim pots should always be turned to their highest resistance when swapping in a new tube.

OPA2134 and its relatives are fairly common opamps for audio. This was a good place to start. The authors would like to know how other FET input opamps perform and welcome feedback from builders. The OPA551, for example, is a FET input opamp that drops right into the Stoopid. However, it only comes in single packages so you will have to account for this with the build.

FET input opamps are preferred because there is a risk that BJT input opamps may tend to excessively load the tube and defeat the effect of the CCS. To use BJT input opamps see the section below for modifications to do this. The authors welcome feedback on the performance of the SOHA with BJT input opamps.

A BUF634 could easily be put into the unity gain feedback loop of the opamp to give super high output. One change that might be necessary if really trying to pull 200mA is to increase the size of the input capacitors in the bipolar PS to more like 2200uF. Even larger values may be required to get full bass.

Mark added 150 Ohm resistors (R6) at the outputs as this enables the amp to drive low and high impedance headphones without experiencing a large change in volume. These can be left out of the circuit at the builder’s discretion, however, their use is recommended. Likewise the pairs of 1N4148 diodes connected to the non-inverting inputs of the opamps are optional, but serve to protect the opamp inputs from overload and their use is recommended.

Here are a few details to pay attention to during construction and double check before applying power to your SOHA:

  • Wiring the Triad transformer is not intuitive; the pins are not numbered consecutively. Study the datasheet carefully.
  • The 78L12 and 79L12 do not share the same pinout.
  • The capacitors in the heater supply (as well as those in the negative half of the bipolar supply) have their positive leads grounded.
  • Use of a star-ground is highly recommended.
  • Use of shielded cable from the input jacks to the pot, from the pot to the tube grids, and from the opamp to the output jack is highly recommended. Attaching the safety ground to the star ground is optional. Most builds have worked fine with the star ground floating but an occasional unit has been quieter with the safety ground connected to the star ground.
  • Grounding the pot body is usually required to eliminate static/hum.

Wire dress is important in this amp to avoid hum. Keep all signal wires away from the transformer; keep the filament wires as far away from the audio circuit as possible.

PC Boards

We’ve created Express PCB boards for the SOHA. These are related to the full schematics with part numbers shown.

For the J113 version eliminate R7, R17 and change R8, R18 to 1k5 1/8W. For the 1N5297 version simply replace the entire CCS with the single diode.

ExpressPCB and PDF files for both the amp and power supply are included below. The tube socket on the amp board is in the center of the board. Note that the tube socket mounts on the foil side of the board. With this configuration you can easily mark a hole in the center of the standoffs and punch it out to pass the tube through so that the tube can stick up through the chassis while the components are sticking downward.

The copper layer in these PDF and ExpressPCB files can be used for home etched boards.

SOHA Amplifier Board (PDF)
SOHA Power Supply Board (PDF)
SOHA Amp and PS boards (ExpressPCB)

Techniques for making home PCBs were suggested by HeadWizer Bill Blair. Here are some links that Bill used to make his own SOHA boards using the single layer PDFs:

EasyPCB Fabrication
HomeBrew Printed Circuit Boards

The boards can be jumpered to use all three versions of the CCSs and to operate as standard plate drive or as source follower drive. This is the stuffing guide for these possible configurations.

cavalli2_fig12
Click here to see full-size stuffing guide.

Figure 6b – Bill’s stuffing guide for the SOHA amplifier PCB

Setup

Wire everything up but don’t put the tube/opamp in yet. Measure the voltages at the B+, V+, V-, and heater. They should be >80V, +12V, -12V, and -12.6V respectively. If they are not then there is a problem that must be fixed before inserting either the tube or the opamp.

If voltages are good and nothing has fried, power down and then insert the tube and opamp. Before powering up again, dial your trim pots so that they are in the maximum resistance position. This will put the maximum bias on the tube. Measure the voltage at the plates (pins 1 & 6) and adjust the associated trim pot until the voltage comes down to 40V for each plate. After these adjustments, measure the B+ again. It should be between 55-65V. Occasionally you may find a NOS tube does not work well in this circuit. You may need to replace the tube to get good results. If so, the tube is probably outside of its published operating characteristics. Each triode of 12AU7/ECC82 draws only 1mA from the B+ supply and at these low currents there can be a wide variation in operating characteristics, particularly among tubes that may be marginally within spec.

Results

OK, how does it sound? Well, in short, stoopidly good. When first powered up, the prototype plate voltage was only 17V and the amp sounded decidedly solid state. Very “steely” and just tonally “off”. As the plate voltage was raised the sound became more lush and tube-like. At 40V the amp began to perform extremely well. The SOHA easily rivals the Cavalli-Jones/Morgan Jones which costs over three times more to build! It’s got decent amounts of bass, classic sweet tube midrange and plenty of top end extension. Also the soundstage is extremely wide and respectably deep. This thing is just plain stoopid fun to listen to!

The amp drives headphones of any impedance between 16 Ohms and 300 Ohms without problems, which covers most that are currently available.

The compression applied by running the 12AU7 with 40V at the plate allows an unexpectedly refined sound with no sharp edges, yet without being mellow. It will reproduce transients when required and has a respectable dynamic range. The overall result is something than can be listened to for extended periods with no “listening fatigue” and providing a pleasingly wide and reasonably deep sound stage.

As is the case with tube amps, a warm up time is required and in this respect the authors agree 20 minutes is required for it to sound its absolute best, but of course it’s up and running after 30 seconds.

Mark has compared this amp to three other headphone amplifier designs available at HeadWize having built them: namely the CJ, the CL MkII, and the BCJ MkI, (the BCJ MkII was unavailable). All of the alternative designs used for comparison tests are more expensive to build, all require potentially lethal voltages, and all are optimized to ensure the tubes are working at their optimum.

Clearly, the SOHA would be the worst of the bunch? Not so. It proved itself to equal the CJ and gets closer than expected to the CL MkII. That’s pretty impressive stuff for a tube amp deliberately designed to be cheap and not use lethal voltages.

Tube rolling in this amp is also a lot of fun. Mark and Bill, who listen mostly through Sennheiser HD-600’s, found that grey-plate 5963’s from GE and RCA and Brimar sounded better than other tubes. Some other Headwizers with low-Z cans seemed to prefer black-plate versions of these tubes. Among the new production tubes, the Electro-Harmonix 12AU7 seemed to approach (but not exceed) the performance of the NOS tubes while the JJ 12AU7 was a somewhat distant second. Differences between tubes seemed to be in the clarity of the top end and the amount and quality of the bass.

Here are some photos of Bill’s SOHA in its final home.

cavalli2_fig7.jpg
Figure 7 – Bill’s SOHA Top Side and Figure 34 – Bill’s SOHA The Guts

Note 1: BJT-Input Opamps

As noted above the SOHA was designed to use FET input opamps. However, to permit opamp rolling, we’ve created two minor variations that permit the use of BJT input opamps.

Bipolar-input opamps like the TSH22IN, NE5532, or NE5534 can substitute for the 2134. But bipolar opamps will have lower input impedance than the FET input opamps. This will increase the loading on the tube and increase the distortion.

One way around the increased loading is to configure the CCS as an active load source follower. This variation requires only one change in wiring at the CCS and will only work for the LND150 and the J113 versions. An active load source follower is a variant of a well-known tube topology where the CCS that is acting as a plate load is also utilized as the output device in a follower configuration. With this topology the output impedance of the gain stage drops considerably and its ability to supply current increases in proportion. With both FET topologies we can wire the FETs as source followers to make a hybrid follower configuration for the first stage.

If you’re using the LND150 CCS you can convert the CCS into a source follower by simply changing the point where the coupling capacitor is connected. The FET then becomes a source follower with low output impedance and the ability to drive higher currents into the load.

cavalli2_fig8.png
Figure 8 – Changing the LND150 CCS for BJT-input Opamps

If you’re using the J113 CCS you can covert it to a source follower using the same technique:

cavalli2_fig9.png
Figure 9 – Changing the J113 CCS for BJT-input Opamps

Although the 1N5297 CRD is actually a JFET wired as a CCS we cannot access the source of the device so the CRD cannot be used when driving BJT opamps.

For example, a full amplifier schematic for the standard LND150 CCS with BJT opamp is:

cavalli2_fig10.png
Figure 10 – Driving BJT input opamps

If your amplifier exhibits high DC offset with BJT opamps, you can decrease the value of R5 from 1M to 100k or even 50k without overloading the gain stage. Note that decreasing the value of R5 while leaving C2 at 100nF also reduces the low frequency response of the amplifier. To correct for this, increase the value of C2 so that the product of R5 x C2 is the same as 1M x 100nF. For example, if you decrease R5 to 100k, then to maintain the same low frequency response, increase C2 to 1uF.

For BJT input opamps, the full schematic is this:

cavalli2_fig11.png
Click here to see full-size schematic.

R1 100k Log Pot C3-C6 100u 100V
R2, R12 300R 1/8W C7, C8, C11 47u 16V
R3, R13 560R 1/8W C9, C10, C12 470u 35V
R4, R14 2k Trimpot D1, D2, D5, D6 1N4148 or similar
R5, R15 100k 1/8W D3, D4 1N4002
R6, R16 150R 1/8W U1, U2 BJT opamp, dual or single
R9 2k2 1/8W V1 12AU7/ECC82 or equivalent
R10 1k3 1/8W BR1, BR2 100V 1A Bridge Rectifier
R11 11k 1/8W VR1 12V Fixed Regulator 78L12
R7, R17 1k 1/8W VR2 -12V Fixed Regulator 79L12
R8, R18 360R 1/8W VR3 Adj. Negative Regulator LM337
C1, C13 1000u 10V T1 30VCT 400mA Transformer
C2, C14 1u 100V

Figure 11 – Full SOHA with BJT opamp, both channels and PS

Note the part changes shown in red. These are the only changes necessary to use BJT opamps in the SOHA. The layout of amp does not change.

Note 2: CCS Comparisons

The choices for standard CCS and acceptable variations are derived from three criteria:

  1. maximum breakdown voltage of the CCS
  2. current regulating ability
  3. PSRR

These comparisons were done using PSpice simulations. These simulations are not likely to give absolute accuracy, but they are good at providing a relative comparison among the various CCSs.

Simulations were done for the following CCS types:

  • Single LND150
  • Single J113
  • Single PN2907A
  • Single 1N5297
  • Single PN2907A with CRD bias string
  • Cascoded J113
  • Cascoded PN2907A
  • Cascoded PN2907A with CRD bias string

This table shows the current variation, PSRR, and breakdown voltage (BV) for these various configurations:

Current Variation

Ripple at Plates

BV

300mVp 1kHz Input

1mVp 120Hz Ripple from PS

Delta I (uV)

Delta V (mV)

PSRR (db)

Topology

Cascoded JFETs (J113)

0.01

0.002

-54

35

Single MOSFET (LND150)

0.9

0.0025

-52

500

Cascoded BJTs with CRD

0.26

0.0095

-40

60

Single BJT w/ CRD

1.2

0.0135

-37

60

CRD (1N5297)

6.6

0.017

-35

100

Single JFET (J113)

13.3

0.034

-29

35

Cascoded BJTS (PN2907A)

0.34

0.058

-25

60

Single BJT (PN2907A)

1.2

0.06

-24

60

The cascoded JFETs have the best current regulation, followed by the MOSFET. The cascoded BJTs with CRD and without CRD have the next best current regulation. This might make these the next best choices. But, we must look at the PSRR and BV tables too.

The cascoded JFETs also have the best PSRR but they have a low BV. The LND150 has nearly the same PSRR (indistinguishable as a simulation result) but a very high BV. The LND150’s current variation comes in fourth behind the cascoded BJTs. However, the PSRR for the cascoded BJTs is 12db and 27db less than the LND150. Furthermore the BV for the BJTs is on the margin of where the voltages in the amp may be, and the BJT CCSs require many more parts than the either the JFETs or the MOSFET.

Taking all of these results together, the LND150 rises to the top for the standard SOHA because of its good current regulation, excellent PSRR, very high BV, and low parts count. The cascoded JFETs come in second because of their excellent regulation, PSRR, and low parts count. The CRD comes in third because of its good regulation, high BV and extreme simplicity (only one part).

Appendix: Simulating the Amplifier in OrCAD PSpice

Alex Cavalli has provided the project files for simulating this amplifier using OrCAD Lite circuit simulation software. The simulations will run in OrCAD Lite 9.1 or 9.2 only (later versions of OrCAD Lite and OrCAD Demo are more restrictive and will not run the simulations). The installation files for OrCAD Lite 9.1 or 9.2 can be downloaded from various educational sites on the internet. Search for them using the keywords OrCAD or Pspice and 9.1 or 9.2. OrCAD 9.1 is the smaller download (27MB). If you have trouble finding these files, email a HeadWize administrator for help.

There are 4 programs in OrCAD Lite suite: Capture, Capture CIS, PSpice and Layout. The minimum installation to run the amplifier simulations is Capture (the schematic drawing program) and PSpice (the circuit simulation program).

Download Simulation Files for SOHA Headphone Amplifier

After downloading cavalli2_soha_sim.zip, create a project directory and unzip the contents of the cavalli2_soha_sim.zip archive into that directory. Move the .lib and .olb files into the \OrcadLite\Capture\Library\PSpice directory. These are the component libraries containing the SPICE models for the vacuum tubes, MOSFETs and opamps used in the SOHA. (Note: heater connections are not required for any of the triode models.) In OrCAD’s Capture program, open the stoopid.opj project file.

The two basic types of simulation included are frequency response (AC sweep) and time domain. The time domain analysis shows the shape of the output waveform and can be used to determine the amplifier’s harmonic distortion. They both run from the same schematic, but the input sources are different. For the frequency response simulation, the audio input is a VAC (AC voltage source). The time domain simulation requires a VSIN (sine wave generator) input. Before running a simulation, make sure that the correct AC source is connected to the amp’s input on the schematic.

cavalli2_sim.png

The following instructions for using the simulation files are not a complete tutorial for OrCAD. The OrCAD HELP files and online manuals include tutorials for those who want to learn more about OrCAD.

Frequency Response (AC Sweep) Analysis

  1. Run OrCAD Capture and open the project file stoopid.opj, if not already open.
  2. In the Project Manager window, expand the “PSPICE Resources|Simulation Profiles” folder. Right click on “Schematic1-ac” and select “Make Active.”
  3. In the Project Manager window, expand the “Design Resources|.\cavalli.dsn|SCHEMATIC1” folder and double click on “PAGE1”.
  4. On the schematic, make sure that the input of the amp is connected to the V4 AC voltage source. If it is connected to V3, drag the connection to V4.
  5. To add the triode library to the Capture: click the Place Part toolbar button (orcad1.gif). The Place Part dialog appears. Click the Add Library button. Navigate to the triode.olb file and click Open. Make sure that the analog.olb and source.olb libraries are also listed in the dialog. Click the Cancel button to close the Place Part dialog.
  6. From the menu, select PSpice|Edit Simulation Profile. The Simulation Settings dialog appears. The settings should be as follows:
      • Analysis Type: AC Sweep/Noise
      AC Sweep Type: Logarithmic (Decade), Start Freq = 10, End Freq = 300K, Points/Decade = 100
  7. To add the triode library to PSpice: Click the “Libraries” tab. Click the Browse button and navigate to the the triode.lib file. Click the Add To Design button. If the nom.lib file is not already listed in the dialog list, add it now. Then close the Simulation Settings dialog.
  8. To display the input and output frequency responses on a single graph, voltage probes must be placed on the input and output points of the schematic. Click the Voltage/Level Marker (orcad2.gif) on the toolbar and place a marker at grid of U6. Place another marker above R9 at the amp’s output.
  9. To run the frequency response simulation, click the Run PSpice button on the toolbar (orcad3.gif). When the simulation finishes, the PSpice graphing window appears. The input and output curves should be in different colors with a key at the bottom of the graph.
  10. The PSpice simulation has computed the bias voltages and currents in the circuit. To see the bias voltages displayed on the schematic, press the Enable Bias Voltage Display toolbar button (orcad5.gif). To see the bias currents displayed on the schematic, press the Enable Bias Current Display toolbar button (orcad6.gif).

Time Domain (Transient) Analysis

  1. On the Capture schematic, make sure that the input of the amp is connected to the V4 sinewave source (VAMPL=0.4, Freq. = 1K, VOFF = 0). If it is connected to V3, drag the connection to V4.
  2. In the Project Manager window, expand the “PSPICE Resources|Simulation Profiles” folder. Right click on “Schematic1-signal” and select “Make Active”
  3. From the menu, select PSpice|Edit Simulation Profile. The Simulation Settings dialog appears. The settings should be as follows:
      • Analysis Type: Time Domain(Transient)
      Transient Options: Run to time = 80ms, Start saving data after = 40ms, Max. step size = 0.001ms
  4. To display the input and output waveforms on a single graph, voltage probes must be placed on the input and output points of the schematic. Click the Voltage/Level Marker (orcad2) on the toolbar and place a marker at grid of U6. Place another marker above R9 at the amp’s output.
  5. To run the time domain simulation, click the Run PSpice button on the toolbar (orcad3). When the simulation finishes, the PSpice graphing window appears. The input and output curves should be in different colors with a key at the bottom of the graph.
  6. To determine the harmonic distortion at 1KHz (the sine wave frequency), harmonics in the output waveform must be separated out through a Fourier Transform. In the PSpice window, press the FFT toolbar button (orcad7.gif). The PSpice graph changes to show the harmonics for the input and output waveforms. The input and output curves should be in different colors with a key at the bottom of the graph.
  7. The fundamental frequency at 1KHz will have the largest spike. The other harmonics are too small to be seen at the default magnification. In the PSpice window, press the Zoom Area toolbar button (orcad8.gif) and drag a small rectangle in the lower left corner of the FFT graph. The graph now displays a magnified view of the selected area. Continue zooming in until the harmonic spikes at 2KHz, 3KHz, etc. are visible.
  8. Harmonic spikes should exist for the output waveform only. The input is an ideal sine wave generator and has no distortion. To calculate total harmonic distortion, add up the spike values (voltages) at frequencies above 1KHz and divide by the voltage at 1KHz (the fundamental).

Note: simulations only approximate the performance of a circuit. The actual performance may vary considerably from the simulation as determined by a number of factors, including the accuracy of the component models, and layout and construction techniques.

c. 2006 Alex CavalliMark Lovell and Bill Pasculle (remove _nospam_).

A Precision Preamplifier-Power Amplifier System with Natural Crossfeed Processing.

by Jan Meier

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“The better is the enemy of the good.”

The headphone amplifier with the natural crossfeed filter published on HeadWize in fact was my first DIY-project. I built this device because I was unsatisfied with the sound reproduction via the headphone-socket of my CD-player. But, as things go, I started to like constructing and decided to design and build some power amplifiers also. Having finished these, they sounded so good that I also made a new matching preamplifier with integrated headphone-amp.

Although the circuit of this new preamp basically is the same as that of the original headphone-amp, some modifications made it possible to increase the sound-quality substantially. The new preamp also has some more options as far as inputs and outputs are concerned. In this article I’ll briefly discuss each modification and leave it to the reader which modifications/options he wants to realize. For the basic preamp/headphone amplifier circuit, the reader is referred to the original article.

The matching 35W stereo power amplifier has 44 output stage opamps per channel and is not intended for a DIY novice. [Editor: the author also includes instructions for building a less ambitious 10W stereo amplifier.] In my opinion it really requires quite a lot of experience to build this amp properly. I had to drill/solder over 2000 holes/connections per amplifier. I made three of them, two for myself for biamping purposes and one for a friend. However, I have found the sound quality of the amplifier to be very rewarding. I was able to compare it with some very decent commercial amplifiers (DENON, LINN, NAIM), but these were completely outclassed by the new preamp-poweramp combo (an opinion shared by others).

THE PREAMPLIFIER-HEADPHONE AMPLIFIER

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Modifications to the original headphone amp circuit:

1. Breaking the ground loop
The preamp incorporates a ground-loop breaker. The ground-connection of the mains-socket is directly connected to the case of the preamp. It is connected to the ground-plane of the audio-circuit via a 4.7 ohm resistor in parallel with a 100 nF capacitor. This resistor prevents 50/60 Hz currents from flowing freely along the ground connections between the various audio components in a system, and thus eliminates the 50/60 Hz hum. Even if one component does not have a loop breaker, but all the other ones have, then there are also no ground loops and there is no problem. The 100 nF provides adequate RF-shielding. To prevent high voltages on the interconnect cables in case of a defective transformer, both inputs of the transformer are secured by a fuse (you never know which input is connected to neutral and which is connected to the alternating high potential).

The metal housing of the mains filter and the enclosure are both directly connected and are grounded to the mains. Normally the “signal ground” is also directly connected to this ground; however, in such situations a ground loop will occur if other equipment is connected to the preamp. By connecting the signal ground through a 4.7 ohm resistor, loop currents (and thereby hum) are greatly reduced. This implies that the preamp audio inputs and outputs MUST have floating grounds – their grounds cannot be directly connected to the enclosure.

2. Driving the opamps into class A operation
The output of each opamp is connected via a 1.5K ohm, 0.6 Watt resistor to one of the voltage-rails to drive the opamps into class A operation. At zero voltage output each output-stage now has to drive a 10 mA current and effectively works in class A. Only driving a low-impedance headphone at high volumes will result in the output stages leaving the class-A range.

By comparison, the output stage of a class B amplifier has two transistors that act like switches. One is opened to deliver the positive output currents, the other is opened to deliver the negative output currents. The switching behaviour going from positive to negative output currents (and vice versa) introduces distortion (for a very short moment the opamp is not able to “control” the signal) in the output that is readily heard (TIM-distortion).

With the output of the opamp connected via a resistor to one of the voltage rails, the DC output voltage will not change but one of the two output transistors will be opened to “dissipate” the current that flows through the resistor. As long as this current is higher then the current demand to drive the load, this output transistor will stay opened (and the other one will stay closed). There is no switching and therefore no distortion added.

This technique in principle does not limit voltage-swing, but it does limit the current swing. However, this should be no problem with my design. I enforce a DC output current of 10 mA. If higher currents are demanded by the circuitry (headphone) driven, the opamp will turn to class AB-operation. It is rather unlikely though that the preamp will need to output currents in excess of 10 mA, and if it does, sound levels will be so high that the distortion will not be heard. This modification resulted in a substantially improvement of sound quality, and can be easily added to the original design. Strongly recommended.

3. RF-shielding and prevention of oscillation
The + input of the first-stage opamps are connected to the potentiometer via two 1.5K ohm resistors. In the middle these two resistors are connected to ground by a 47 pF capacitor. Also 10 pF capacitors are added between the outputs and the inverting inputs of each opamp. These measures prevent high-frequency signals from entering the circuit and thereby increase stability and prevent high-frequency oscillations. I used polystryrol capacitors, but any other film-capacitors will also do.

4. Bass-enhancement circuit

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I slightly modified the bass-enhancement circuit. The functionality has not changed, but now the feedback resistors are 10K ohms, and the outputs of the opamps are always connected by a 150 nF capacitor. This does not improve sound quality, but it does prevent annoying clicks when changing the settings of the bass-enhancement.

5. Decreased impedance of the potentiometer
Originally, a 50K ohm potentiometer was used. I found a lower impedance to sound marginally better – but only marginally. It is not worthwhile replacing a 50K ohms pot, if you already built the original circuit.

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Additions to the original headphone amp circuit:

1. Inputs
There are now five inputs connected via a switch to the potentiometer. At the input jacks, the signal-pathways are connected to ground by 47K ohm resistors. This decreased the capacitive and inductive crosstalk between the various channels/inputs both audibly and measurably. Actually, I was rather surprised how much these resistors added to the sound quality.

2. Line out
For recording purposes, a non-volume-controlled output was added. The audio source can be chosen independently from the source being listened to. Note that there is no signal buffer and that it might be advantageous not to have these switches set to the same position, if a recording device is connected. Otherwise, the same source will be loaded by both the preamp and the recording device and cables.

3. Preamp out
A volume controlled output to drive a power amplifier. This output signal is not processed by the natural crossfeed filter.

4. Processor out
A volume controlled output to drive an amplifier (e.g., an electrostatic headphone amplifier). This audio-signal is processed by the natural crossfeed filter.

5. Headphone out
For connecting a dynamic headphone. The headphone jack I used has a built-in switch that disconnects the processor outputs, if a headphone is connected. It is made by Lumberg (part-number is KLBRSS 3 L) and can be ordered at Farnell in Germany (ordering number 838 550). The jack is directly mounted to the board.

6. Increased headphone output impedance
The headphone output impedance is normally near zero ohms. Optionally, the output impedance can be increased to 120 ohm by adding a resistor. Many headphones are designed to be connected to a source with a 120 ohm output impedance. Personally, I did not add these resistors to my preamp, but built a plug to connect preamp and headphone that has these resistors incorporated. My Sony headphones reacted very favorably to this increased impedance, whereas my Sennheiser HD600 became rather muddy. Simply try which suits your headphones/taste best. Since most dynamic headphones have a higher impedance at lower frequencies the increased output impedance results in an increased bass (with my Sony + 3dB!, Sennheiser + 1.5 dB).

THE POWER AMPLIFIER

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The design

This all-opamp power amplifier has 44 output opamps per channel! Why 44 output opamps? My design goals were:

  • The output stage should be very fast.
  • The output stage should be linear, so the “control-opamps” would have an easy job.
  • The amplifier should be driven by a regulated power supply (unregulated supplies, as used in conventional amplifiers are IMHO a major source for a decrease in sound quality, because there is no infinite PSRR).

I wanted a completely regulated power supply for the output stage for currents up to 4 Amps. This implied using 4 pairs of LM317/LM337, since one voltage regulator only handles 1 Amp. I, therefore, would also need at least 4 pairs of output transistors per channel, since you can’t put voltage regulators in parallel to supply the same component with current. (There are voltage regulators that handle more than 1 Amp, but these are very expensive and require lots of heatsinking). So the choice was between [8 transistors + 4 opamps + heatsinks] or 44 opamps.

I also wanted to have a linear output stage with no distortion, which implied local feedback using one opamp per transistor-pair. For semi-class-A operation, 4 pairs of transistors allow for 4 different “switching” points (or more precisely, the output voltage where the opamp switches between the two transistors in the output stage). 44 Opamps allow for 44 different “switching points” (actually I only use 24 different points but this still is far better than 4). Each transistor-pair that is to be driven in class-AB dissipates heat and requires a heat sink. Opamps like the LM6171 don’t need a heatsink (unless you use the dual version at high currents).

Opamps are an ideal solution, but their current capabilities are too limited. I, therefore, placed 44 of them in parallel. To drive them in pure class A would demand a high DC-current (per opamp) and increase power dissipation. I decided to inject only a relatively small DC-current, so each opamp works in class AB.

By using different current values for the various opamps, each opamp will switch at a different overall current demand. At any time of operation, the major part of the output opamps will be in a non-switching state, and the “control-opamp” (which is working in class-A) is able to control the output signal continously. TIM-distortion is eliminated, although class-AB functionality is used. In contrast to a conventional class AB device, where there is no control during switching, there always will be “control” using many parallel output stages, each switching at different points. That’s why I called it semi-class A.

I wanted biasing currents for the output stage opamps between approximately 1.5 and 5 mA – not too high to cause excessive current drain and power dissipation, and not too low to start switching at very low sound levels. So RP resistors should be approximately within the range 3K ohms to 12K ohms. Then I simply selected values that were available in the catalogue. No sophisticated calculations.

When you look at the costs, I don’t think that the transistor solution is much cheaper than the all-opamp solution. High quality, high speed transistors as well as decent heatsinks do cost. Of course the second solution is more elaborate, but the way is our goal, so time is for free.

Construction

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There are no special new construction techniques. This is not a project for novices. The enclosure shown is a real nice part but unfortunately also very expensive ($150). Like the preamp, the audio input and output jacks MUST have floating grounds, because of the loop-breaker circuit in the power supply.

The opamps that have the highest power dissipation in “standby” are the input opamps. They have to dissipate a current of 18V/1.5K ohms = 12 mA. Power dissipation is 216 mW. This is fully acceptable. Power dissipation of the output stage opamps is much lower, except when driving large signals into low loads. However, such power demands are transient. Power handling is one of the reasons I decided to use the LM6171 instead of its dual version LM6172. Note that 2 x 44 amps have quite a large total body surface so any heat is easily transferred to the air.

Originally, part of the amp got quite hot, not because of the heat dissipated in the opamps, but because of the heat dissipated in the voltage regulators. The total standby current for both channels is approximately 0.34 A. The power transformers are 2 x 18 VAC (50 Watts). With a voltage drop across the positive voltage regulators of 25V – 18V = 7V, power dissipation becomes 2.4 Watts. The negative regulators have to dissipate an equal amount of heat, so total power dissipation in the voltage regulators comes up to approximately 5 Watts. This is easily handled by the heatsink I made out of a aluminum sheeting. Even at high sound levels, the output voltage of the regulators does not drop below 23V, which means that the regulators can still do their jobs most adequately.

The amplifier uses 2 power transformers, not to have a completely independent supply for both channels (because they aren’t), but because I use a very slim enclosure and one big transformer would not fit. Each transformer “drives” 4 positive and 4 negative voltage regulators (4 pairs). Each voltage regulator pair, consisting of a positive and a negative voltage regulator, “drives” a group of 11 output stage opamps (44 opamps total). One additional pair of positive and negative voltage regulators (after a thorough LC-filtering) powers the the four input stage opamps of both channels. There are separate fuses for both transformers. The values of the fuses shown in the schematic are for 230VAC mains. For 110VAC, a value of 800 mA would be more appropriate.

A simpler 10W amplifier

The maximum output voltage of the amplifier is approximately 16V and the maximum current is about 6.6 Amps. To build a “smaller” power amplifier, reduce the number of output stage opamps to limit the current capability of the output stage. There will be a point where the amplifier will not be able to deliver the 16V. It all depends on the impedance of the speaker. For example, using 20 opamps will limit the maximum output current to approximately 20 x 0.15 = 3 Amps. With a 4 ohm loudspeaker, the maximum voltage is 12 V and maximum music power is 0.5 * I * I * R = 18 Watts per channel.

Which output stage opamps should be removed to reduce the output power? I would take the ones with the higher impedances to the power rails first, since this would drive the output stage for a longer period of time in pure class A. However, I think the sonic difference will be small, if some of the other opamps were removed.

Operation

The maximum power of the amplifier is primarily set by the supply voltage (18V). The maximum output voltage is approximately 16V. With 8 ohm speakers, the power rating is 16 Watts per channel. With 4 ohm speakers, the power rating doubles (32W per channel). Continuous power output equals the peak power output since, except for the supply voltage, power supply is “over-dimensioned”. Each LM6171 opamp is able to deliver up to 150 mA of current, we have 6.6 Amps per channel. With 16V output, the amp should drive loudspeakers down to 2.5 ohm. In this case, the output impedance effectively seen by each separate opamp is 44 x 2.5 + 10 = 120 ohms and does not represent a major problem (LM6171 is specified for impedances down to 50 ohms).

The noise of preamp and power amp measured at the loudspeaker connections with the volume at maximum was heardly noticable (no hum due to the regulated power supply) and unmeasurable for my multimeter (less then 0.1 mV!). SNR thus by far exceeds the specifications of the CD and is estimated to be better than 120 dB.

The “Phase” switch can be used to configure the both channels of the amplifier for biamping or can convert the amplifier into a monoblock with double the output power. It has three positions:

  • Position 1: each channel is driven by its own input buffer. This is the normal stereo mode.
  • Position 2: each channel is driven by one and the same input buffer. This can be used for biamping when both channels drive different units of the same loudspeaker. (Alternatively, one can connect the same output of the preamp to both inputs, but this solution saves cable and was given for free by the phase-switch.
  • Position 3: Each channel is driven by the same input buffer but the phase of one of the output channels is reversed. Connecting one single loudspeaker to the positive terminals of both channels (instead to one positive and one negative (ground) terminal) doubles the signal amplitude. This option is specially designed for high impedance, low efficiency loudspeakers. Advantage: Maximum output power per loudspeaker has increased by a factor 4 (approximately 64 Watts at 8 ohms instead of 16 Watts). Disadvantage: The stereo amp is converted to a mono amp (double costs).

Last week I listened to some real loud music. Due to the Analoguer filter (described in another article on HeadWize), I am simply able to sustain much louder sound levels now. I have used the amps with various loudspeakers (CHORD, KEF, QUAD, etc.) and although some of these speakers are quite hard to drive, the amp did not seem to have any problems with them. Especially the excellent bass-control was one of the first characteristics noticed by most listeners.

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I am offering a DIY-kit for the headphone amplifier (NOT the preamplifier) with the updates discussed in this article. The completed headphone amp is shown above. If you are interested in the kit, please e-mail me.

I strongly recommend experimenting with these designs. As always, have fun!

Addendum

10/11/2000: Corrected ground-loop breaker section in power supply schematics for preamp and power amp. For the ground-loop breaker to work properly, the circuit ground must be isolated from the metal enclosure, which is connected to the mains ground.

11/6/2000: Repositioned 10pF feedback capacitor around IC2 in preamp for greater stability. Also added pictures of headphone amplifier kit and updated picture of preamp-power amp combo at beginning of article.

c. 2002 Jan Meier.

A Compact 50W Integrated Amplifier with Meier Headphone Section.

by Tim Harrison

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My reason for constructing this project was to develop a design for a compact integrated stereo amplifier suitable for use by a poor (but sound quality conscious!) student living in a university or college dorm. The amplifier drives a pair of loudspeakers using two LM3876 integrated power amp ICs (50 watts per channel), or a pair of headphones via a Meier crossfeed filter and an OPA2134 dual opamp. It provides four switchable line level inputs, and an unbuffered line level output for recording purposes. The design uses readily available good quality components, and is based around four separate PCBs; one for each power amp channel, one for the power supply board, and one for the preamp/headphone driver.

THE CIRCUIT

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Figure 1

The block schematic for one channel of the design is shown above (figure 1). The preamp and the first stage of the headphone amp are separate in this application, ‘straddling’ the gain across the volume control. There is an initial gain of 2.5 before the control, followed by a further gain stage of x3 after it. This arrangement allows the power amp to be driven directly from the output of the volume control without further gain, and makes for lower noise operation of the headphones. The input selector switch is a 4-way, 3-gang type, so one gang isvused for each channel, and one gang is used to switch the input indicator LEDs.

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Figure 2

Above is the schematic for part of the preamp board (figure 2). The output of the selector switch is sent to pins J1 and J3. Looking at the left channel, C1 and R2 form a low pass filter with a -3dB point of 40kHz, which rejects any RF interference picked up on the interconnects. R2 also sets the input impedance of the unit, in this case 47k ohms. R1 ensures the opamp U1 is presented with an equal impedance at both its inputs, helping improve its distortion performance as outlined on the OPA2134 datasheet. The value of R1 (9k1) is the nearest commonly available value to the parallel combination of R3 and R4 (22k and 15k respectively). R3 and R4 set the gain of this stage, just under 2.5 in this case. This value allows ample headroom for a wide range of source signals, which could be as much as 3VRMS. In this case, the peak output voltage of 10.6V would be fine with the suggested ±15V power supply.

This initial gain brings the signal up to a level whereby the output from the volume control can drive the power amp circuits directly, with no further gain, and allows the headphone driver circuit to operate with a lower gain, giving lower noise performance. C7 forms a 100kHz low pass filter with R3, rolling off the gain to unity at very high frequencies, and helping promote stability of the opamp. It is not strictly necessary with the suggested OPA2134 device, but allows the drop-in substitution of a cheaper but more oscillation prone device, such as the NE5532, if budgets are tight. C19 AC couples the output from this stage to the volume control, and with a 50k potentiometer, sets the -3dB point of the headphone amp’s response at 1.4Hz (the power amp has further high pass filtering). This capacitor is very important, as all the other stages are DC coupled, and C19 prevents any DC offsets from source components being amplified and presented to the headphones or speakers.

The resistor R9 links the output of the input selector to a recording device, such as a tape deck or minidisc recorder. It helps prevent the source becoming too loaded down feeding both the input gain stage and the recording device, and protects the source should the output become shorted to ground for any reason. The outputs from J5 and J6 are fed into the volume control pot, which should be a good quality type. Finally, C3 to C6 provide local decoupling of both the power supply rails, C5 and C6 decoupling the high frequencies, with C3 and C4 decoupling the lower ones.

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Figure 3

The output of the pot feeds the power amp and the headphone driver, which is also mounted on the preamp board. Looking at the above schematic for the headphone driver (figure 3), we can see that the opamp U2 is used in a similar configuration to the input amp U1. In this case, R24 matches as closely as possible the parallel combination of R11 and R12, helping reduce distortion as before. Again, C21 allows compatibility with cheaper opamps. R11 and R12 set the gain of the stage at just over 3, bringing the signal up to a level sufficient to drive a pair of headphones. This stage also acts as a buffer, isolating the Meier crossfeed filter from the varying output impedance of the volume control. C8, R14, (with C10, R21, and R15) form a crossfeed filter, which in this case is permanently wired in circuit. A detailed description of the operation of this circuit can be found in Jan Meier’s article A DIY Headphone Amplifier with Natural Crossfeed.

Basically, the circuit performs a frequency selective mix of the two channels into each other, allowing recordings meant for speaker listening to sound natural on headphones. I had built projects with the filter made switchable in the past, but I never turned it off, so the switch was omitted here. Finally, the opamp U3 forms a simple noninverting buffer to drive the headphones. R17 forms a minimum load when the phones are disconnected, and helps prevent pops and clicks when they are connected with the unit powered up. While it is possible to substitute cheaper opamps in other parts of the circuit, the device used here needs to have a high output current capacity, and must remain stable when driving difficult loads. J10 and J12 are the output to the headphone socket, which should have its ground isolated from the chassis so as not to defeat the ground loop breaker circuit. Again, C11 to C18 provide local supply decoupling for the opamps.

You can find more information on the detailed operation of opamp based circuits, such as the preamp and headphone amp circuits presented here, in Chu Moy’s article Designing an Opamp Headphone Amplifier. Figure 4 is the power amp schematic for one channel (both channels are identical – and use one power amp board each).

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Figure 4

The circuit given here is similar to the one presented in the article, Single Chip 50 Watt / 8 Ohm Power Amplifier, on Rod Elliott’s site, ESP. The LM3876 is a good quality component capable of delivering 56W continuously into an 8 ohm load and 100W peak – enough for any dorm! It has a quoted distortion figure of 0.06% at 40W output, and offers good sound quality in a simple design. It has comprehensive output protection circuitry, preventing not only thermal runaway, but protecting the device from short circuits on the output, and voltage spikes from inductive loads.

Looking at the circuit, R3 and R1 set the gain of the power opamp at 23, and C1 limits the DC gain to unity. It also forms a low pass filter with a -3dB point of 7.2Hz. R2 draws roughly 1.5mA from pin 8, disabling the internal muting function of the LM3876, and C2 provides a large time constant for the action of the muting circuit. R4 should be a 1W resistor, and has 10 turns of 0.4mm enamelled wire wound round it, with its ends soldered to the resistor leads, giving a roughly 0.7uH inductor in series with the 10 ohm resistance. The inductor acts to promote stability of the power opamp, by ensuring a minimum 10 ohm load at higher frequencies. Likewise, the low pass zobel network formed by C7 and R5 (which should also be a 1W type), helps prevent oscillation should any RF appear on the output. C3 to C6 provide local supply decoupling for the power amp IC.

To enable the power amplifier to deliver its full rated power (56W/ch) continuously, and to cater for the potential 100W peaks, I decided to build a good quality power supply for the project, capable of supplying 200W. The main power supply for the speaker amps was built directly into the chassis, and is a fairly standard design. It supplies ±35V, and is capable of just over 3A continuous per rail for both the power amps. A ±15V supply for the preamp and headphone driver is provided from the main supply by the PSU board. Firstly, I will describe the main power supply, whose schematic is shown below (figure 5):

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Figure 5

The mains enters the chassis via a filtered IEC inlet, and the live line is fed through a 1A antisurge type fuse mounted in an insulated chassis fuse holder, before both the live and neutral lines are fed to a DPST rocker switch mounted on the front panel. The mains feed from the switch is connected to the primary of the power transformer, and a pair of transient suppressors are wired in parallel with it (only one is shown in the diagram). They should be rated for the mains voltage where you are, and should be mounted securely on the base of the chassis, I used two sections from an insulated terminal block.

The secondaries of the transformer are wired in series, and the wires from the toroidal types can be connected directly to a heavy duty chassis mounted bridge rectifier. The output of the bridge rectifier is sent to a pair of reservoir capacitors, C2 and C3, connected in parallel with C4 and C5, which provide high frequency decoupling. The only other point about the power supply that needs explaining is the ground loop breaker circuit. The 0V rail is connected to chassis ground and mains earth via R1, a 10 ohm wire wound resistor, in parallel with C1, a mains rated 100nF capacitor. The resistor prevents any currents flowing round the loop created by the mains earth and the ground in unbalanced phono interconnects. The 100nF capacitor shorts the resistor at high frequencies, allowing any RF to flow to ground in the normal way. I placed C1 and R1 on the underside of the stripboard I used to mount the reservoir and decoupling capacitors.

The output from the main PSU is fed to the power amp boards via a front panel DPST switch, allowing the speaker amps to be switched off for headphone only listening, and also (unswitched) to the preamp PSU board. Below is the schematic for the preamp PSU board (figure 6):

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Figure 6

The ±35V rails must be reduced to around ±20-22V before they can be fed to standard three pin regulators. I simply used a potential divider comprising R3 and R6 for the positive rail, and R4 and R7 for the negative rail. Simply placing a reverse biased 12-15V zener diode in series with the supply, i.e. in place of R3 and R4 (and omitting R6 and R7), would be an alternative option, and probably simpler – this option didn’t occur to me until after I built the prototype! C1 to C4 decouple the output of the regulator, and R1, R2, and R5 set the current flowing though the LED indicators, around 15mA in this case. The stabilised ±15V supply is presented on pins J1-J3, and the remainder of the pins provide supplies for a pair of power on indicator LEDs (mounted next to the mains rocker switch), and the input selector LEDs. These are mounted above the input selector switch, and light to show which input has been selected. They are controlled by the remaining gang of the three gang rotary switch.

CONSTRUCTION

I have provided my PCB artwork for you to use to make your own PCBs if you are interested in building all or just part of this project. Below are links to the artwork files, and the relevant placement guides (all GIF format):

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Download PC Board Artwork and Placement Guides 1
Download PC Board Artwork and Placement Guides 2
Download PC Board Artwork and Placement Guides 3
Download PC Board Artwork and Placement Guides 4
Download PC Board Artwork and Placement Guides 5
Download PC Board Artwork and Placement Guides 6

The artwork will print the correct size if you set your graphics software to output 600dpi to your printer. The placement guides should be printed at 300dpi. If you have trouble getting them the right size, the power amp boards should be 41mm wide, the PSU board should be 113mm wide, and the preamp board 132mm. I made the artwork for the lead pitch and size of the components I could source, so I suggest you print out the placement guides real-size (300dpi), and compare the sizes and lead pitches of the components you can source, selecting the ones that best fit the board.

As a guide to component selection, I used 0.6W metal film resistors throughout, except for R3, R4, R6, and R7 in the PSU, which should be 5W wire wound radial lead types, and R4 and R5 in the power amp, which should be 1W wire wound types. For the decoupling capacitors use ceramic disc types, and for capacitors in the signal path (C19, C20, C8 and C10 in the preamp), I used the Wima MKS4 250V series, although any metal film type will do (but may not fit).

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The line level signal from the sources is received via an array of gold plated phono plugs on the rear of the unit. The plugs I used had red and black identification bands on them to indicate which channel should be connected to them. This was important, as I was not planning to print any lettering onto the case, so the connections and controls had to be fairly self-explanatory. The phono plugs should be mounted using an insulating bush, as the design uses a ground loop breaker circuit, and the signal and earth (chassis) grounds are separated.

The source signals are routed via screened cable to a rotary selector switch mounted on the front panel which is used to select the source to be listened to (and recorded from). The switch should be a good quality part, as a positive tactile response from it enhances the feel of the finished project. The part I used was a 3-gang 4-way type, allowing 4 stereo inputs to be accommodated, leaving one gang free to switch the source indicator LEDs mounted above the control. I used a cheap part by a company called Alpha, their SR2611 series. This switch works fine and only cost a few pounds (roughly $5 US).

For a volume control, I used a 50k ALPS pot, but a cheaper type of any value between 10-100k could be used. A conductive plastic track type is preferable to a carbon track, and should be logarithmic law (also called audio taper). The ALPS RK27 series pots (the blue ones), while pricey, come highly recommended, as they have a very nice tactile feel to them, and exhibit good tracking between the gangs.

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For the preamp and headphone section opamps, I recommend the OPA2134 by Burr Brown, and DIL sockets are a good idea to help prevent heat/static damage during soldering. Note, the LM3876T power opamp in figure 4 must be used with my PCBs, the T suffix denotes the package type. The power opamps share a large 2 degrees C per watt heatsink mounted on the rear panel in the prototype and are mounted using greaseless silicone insulators and insulating bushes. Make sure the metal tab of both the power amp ICs is isolated from the chassis – this is very important.

Power supply

The value of the mains fuse in figure 4 varies depending on what type of transformer you use, and the supply voltage in your country. Since I live in the UK where the mains supply is 230V, and I am using a 225VA rated toroidal transformer, a 1A antisurge fuse was used. Take care to get this value right, as if it is too low, you will suffer nuisance blowing, and if it is too high, you will not get proper protection in the event of a fault. The fuse rating can be calculated in the normal way using I = P / V. A double pole type switch is preferable to a single pole type, as it allows the unit to be completely isolated from the mains when it is switched off. The mains rocker switch used should be rated to handle the in-rush current of the transformer, anything over 4-5A should be fine in this case.

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I don’t normally include transient suppressors in the power supplies of audio projects I build, as I run the mains supply to my home audio system through a filter which includes them. However, as this integrated amp was designed to be used away from home, they are included here. You can use a pair together in parallel as suggested to increase their dissipation capacity.

The toroidal power transformer was made by Nuvotem and sourced from RS components in the UK, at a cost of about 23UKP (or about $35 US). In general, the power transformer itself should be a good quality type, and I recommend a 225VA toroidal part for this project. If the budget is tight, a lower value toroidal (say, 160VA) could be used, or even a conventional EI laminate type. Although the standard EI transformers are cheaper than toroidal types, and exhibit a lower inrush current, they are less than ideal for a compact unit. They tend to be quite bulky, and emit strong electromagnetic fields, leading to hum pickup in adjacent circuitry. A toroidal transformer is both compact, and emits a far less strong field.

Although the rated current of the power supply is only 3A, the charging current of the reservoir capacitors will be much higher than this at times. I recommend using a 35A type bridge rectifier, such as the KBPC3506. A pair of heavy duty insulated terminal blocks should be mounted nearby, and the centre tap of the secondaries connected to this. The terminal block will now form a star grounding point, and should be the place all the 0V rails in the unit are connected together. This method of grounding ensures hum free operation. Save yourself a lot of grief and use this method the first time – hum free results are almost guaranteed.

If the budget is tight, 4,700uF capacitors can be substituted for the 10,000uF ones specified in figure 4, especially if a 160VA transformer is used. I had trouble fitting 10,000uF caps into the chassis, so I used two 4,700uF caps in parallel per rail. I couldn’t get any capacitor mounting brackets, so I simply soldered C2 – C5 onto a small piece of stripboard. You could use either method, but be sure to take your DC output from the capacitors and not the rectifier.

For the 5W resistors in figure 5, I used vertically mounting ceramic, wire wound resistors, but you could use standard axial types, with one leg bent down the side, if you find the radial types hard to get. C5 to C8 decouple and stabilise the output of the potential divider (or zener diode), before it is fed into a pair of standard voltage regulators. These should be mounted with a pair of small flag type (clip-on) heat sinks with a thermal resistance of around 20-25 deg. C per watt. I used Redpoint Theramalloy PF752.

Chassis

I mounted the project in a compact instrument case (300mm W x 150mm D x 100mm H), which has a removable internal chassis. The case was supplied painted grey, but once I had drilled it, I decided to repaint the chassis blue to make the project look more individual. I prepared the chassis by sanding it down thoroughly, making sure that all surfaces would provide a good key the paint could adhere to. I then cleaned all the surfaces with white spirit, and applied three thin coats of standard car spray paint. I used diffused blue 3mm LEDs, black rocker switches with blue markings, and black aluminium knobs to complete the effect.

You can see the layout I used pretty clearly from the pics inside the unit, all the boards were mounted on the base of the chassis, except for the two power amp boards which were mounted on the rear panel. The bridge rectifier is bolted to the bottom of the metal chassis. I used 4mm binding posts for the speaker terminals, two black ones for the ground connection, a green one for the left channel, and red for the right. All the signal wiring should be done using shielded cable, with the screen grounded at one end only. Ribbon cable can be used for LED wiring, 32/0.2mm hookup wire should be used for power amp supply and speaker connections, and 7/0.2mm hookup wire can be used for other low power connections.

If there is hum on the output of the completed project, the problem is almost certainly to do with the ground scheme used. Make sure that there is a 10 ohm resistance between the chassis and signal ground (i.e. that you have not defeated the ground loop breaker), and make sure you have not accidentally grounded a point by two paths simultaneously. The star grounding scheme as outlined earlier is highly recommended. The path to ground on the volume control pot is particularly critical, in the prototype the unit refused to stop humming until the far end of the wiper had its own separate connection to the star ground point. It should be possible to set the volume control to zero and, with the unit on, put your ear to the speaker and hear nothing but a faint hiss.

RESULTS

My impression of the project overall is very good, it sounds good, and is very compact. The performance from the IC power opamp is impressive, and I think my prototype looks nice, too! Listen to your favourite cans through it late into the night, or let it provide some serious slam through speakers for a small room or dorm.

c. 2002 Tim Harrison.

A Current-Domain Electrostatic Amplifier for Stax Omega II Headphones.

by Kevin Gilmore

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I bought the Omega II headphones without the amplifier ($1995 + shipping from EIFL Corporation in Japan). I would love to listen to the SRM-007t or SRM-717 amplifier, but really do not want to fork over $4000 to do so. I have been working on this solid state Stax headphone driver for a long time. It satisfies all of the design requirements. Of course it sounds absolutely amazing which is clearly the goal here. There are no capacitors in the signal path. Its fully DC coupled. No expensive parts, and can be built by just about anyone.

The amplifier operates primarily in the current domain. The first stage is a voltage controlled current sink. The second stage is a current-controlled voltage source. The fourth stage is a constant current sink. The main advantage of current domain amplifiers is speed. Standard voltage gain amplifiers with lots of gain are affected by the Miller Effect which prohibits extended frequency response.

This solid state amp is so much better than my tube amp that I no longer listen to it. I’m not a solid state snob; it’s just plain better. The people who have listened to this amplifier (some of whom were giants in the industry in their day) love it, much more than my tube amp. I love it too. I can’t stop listening to it. The tube amp has moved into a secondary position in my listening rack.

How It Works

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The first stage is a differential amplifier with feedback directly from the output stage. It works equally well with both balanced and unbalanced audio input sources. The step attenuators from Goldpoint make good volume controls for this stage. The JFET device is a dual JFET all on one wafer. It is known for extremely low noise and excellent matching, and is used in a number of expensive designs, such as the Nelson Pass amplifiers.

Because the amp is totally DC coupled from input to output, drift in the input stage is a bad idea. Since the first two stages run in current mode, the JFET input is more linear than a pair of bipolar transistors. Dual transistors all on one wafer suitable for audio use are hard to find these days.

The approximate voltage gain of this stage is 5. But it really runs in current mode. The unit was designed to work equally well in both balanced and unbalanced mode. For single-ended signals, ground either the + or – input and apply signal to the other. The much higher impedance of the JFET works better when one side is grounded for unbalanced inputs.

The second stage starts with a constant current source. The current source feeds a common base amplifier. The common base amplifier feeds a modified Vbe multiplier. I believe a famous designer is now calling this circuit a current tunnel. Its the most linear way of translating the voltage down to the bottom rail. The voltage gain of this section is about 4. The basic idea of the first two stages is to supply the third stage with a very fast low impedance drive signal that is referenced to the bottom rail.

The current sources in the second stage supply 2 mA each. With no signal, the FETs take 1 mA, leaving 1 mA going through the common base amplifier into the bottom transistor (which is wired as a vbe multiplier). This generates the 13 volts (referenced to – rail) necessary to properly bias the 3rd stage. The bottom transistor acts like a zener diode in series with a resistor, except a lot less noisy.

The third stage is another differential amplifier feeding another common base amplifier. The simple differential amplifier has a voltage gain of about 100. The common base amplifiers are used to reduce the miller effect on the differential pair. Since the miller effect depends on both gain and output voltage swing, reducing the output voltage swing of the bottom differential transistors significantly improves the speed of this circuit.

The fourth stage is an emitter follower driven by a constant current source (gain = 0.99). This output stage dissipates 12 watts total (3 watts per transistor x 4 transistors). The main design goal was low output impedance. For example, my electrostatic tube amp has a 50K load resistor and thus has a 50k output impedance. This amp has a 25 ohm output impedance (actually a little less with feedback) The result is a much more extended high end. The slew rate of the solid state amp is more than 5 times that of the tube amp.

For the output stage, each 2SC3675 sources or sinks 9 mA at a quiescent output voltage of zero volts referenced to ground. For the driver stage, each 2SC3675 sinks 1.1 mA, resulting in 1 VDC at the collector (referenced to ground). The bases of the 2SC2705s sit at about 16 volts (referenced to – rail). The overall open loop gain of the amplifier is about 2000, but feedback reduces it to 1000. Even without any feedback of any kind the total harmonic distortion of the amp is still under .02%.

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My first prototype, the unit in the pictures, uses an unregulated power supply. Given the stiffness of the capacitors, and the fact that the amplifier is pure class A, there is absolutely no fluctuation in voltage when signal is supplied. Of course, a regulated supply is always better. A regulated design is shown above. The 2SC3675 and 2SA1968 are mounted on heatsinks (the small tab ones are fine). The transformer is a Thordarson 24R22U (Allied # 704-0952). Adjust the pot to get 580VDC for the bias voltage.

The ±15 volt supply is an encapsulated fully regulated power supply brick from Sola Linear (Allied part number 921-9215), which retails for $117. I used a 60 mA version, but thats overkill, because the total current drain is about 12 mA for both channels. Lots of companies make these. It’s the black brick in the picture. It is NOT a switching supply. I do not use switchers in audio stuff if I can possibly help it.

Construction

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Download full size PC board designs and component layout 1
Download full size PC board designs and component layout 2
Download full size PC board designs and component layout 3

This project involves working with high voltages, so be extremely careful! Keep one hand behind your back at all times. 600VDC across both arms might possibly stop your heart.

All resistors are 0.5W. Most do not need to be. The 300K resistors in the top of the 3rd stage need to be 0.5W. The 150K resistor in the current drive in the last stage needs to be 0.5W. I am trying to find 2SA1968 transistors, which are 900 volt PNP types. If they are fast enough, then the two 300k resistors can be replaced with current sources instead, making the amp 100% current source driven.

The LEDs in the amplifier circuit are voltage references (1.7 volts types in the prototype) which track changes transistor voltage with temperature (low voltage zener diodes have tracking problems). They also serve to show that the unit is running properly. If the LEDs are not lit, something is wrong. You could always replace each LED with 3 1N914 diodes in series, but the LEDs look so pretty (reminds me of the glow of a vacuum tube).

I am using standard regular brightness red LEDs. The blue and green ones run at different voltages (blue = 2.6 volts, green = 2.1 volts). Using LEDs with voltage drops greater than 1.7V can affect biasing. Higher LED voltage drops in the first and second stages will tend to cancel each other out, and the numbers will be the same. That is, a higher voltage diode will increase the current sources from 2mA to maybe 3 mA (each), but at the same time, the current sink in the first stage will go from 2mA to 3 mA (total), so the net result is zero.

However in the final stage, a higher voltage diode will increase the standing power. As long as the heatsinking is good, an increase from 12 watts per channel to 15 or so is just fine. The transistors are actually good for 10 watts each, so it is possible to increase the bias to 40 watts per channel.

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All 4 output transistors are mounted on one aluminum angle that bolts through the front panel to the heatsink. The mounting heatsink is 4″ x 5″ x 1/8″ aluminum plate, punched and then bent along the short axis. There are 4 holes that hold the transistors to the angle, and 5 holes that bolt the angle to the heatsink. The blue-finned heatsinks I found on some old power supplies. I used them because they were big enough and pretty at the same time. The 2 2SC3675 drivers have small standup heatsinks.

The two pots balance the output voltages to 0V referenced to ground. Begin the adjustment by putting a voltmeter between + output and – output and setting the first pot for zero volts. Then put a voltmeter between the + output and ground, and set the second pot for 0V. After the amplifier warms up for 30 minutes, adjust the pots again. I adjusted my unit once, and keep checking it every so often. The output voltages on my unit are less than ±200mV. Compared to the 580 volt bias, that is close enough to 0V. And that is over a 1-month period.

Assemble the output stage with care. The full output voltage swing exists between the bases and the collectors of the bottom output transistors. Poor soldering techniques combined with excess flux can cause an arc which may damage the transistors. It happened to me once.

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The Stax jack is Allied part number 719-4043. For all headphones except the Stax Omegas, the plug fits in all the way. On the Omegas, the plug is a little fatter and does not fit in all the way, because the plastic center of the jack is about 0.25″ below the base of the metal rim. So I put the jack in a lathe, and took 0.25″ off the metal rim so that it is flush with the plastic insert. This modification does not affect the fit of other Stax headphone plugs. For details on how to wire the jack, see All-Triode Direct-Drive Tube Amps for Electrostatic and Electret Headphones.

The 2SC3675 is made by Sanyo. The 2SA1968 and 2SA1156 are from NEC. The rest of the transistors are from Toshiba. Here are the current prices:

2SK389 1.90 each
2SC1815 0.30 each
2SC380 .37 each
2SC2240 .55 each
2SA970 .79 each
2SA1156 .82 each
2SC2705 .49 each
2SC3675 1.56 each

In the USA, all of the Japanese semiconductors are available from B&D; Enterprises. B&D; takes credit cards. The entire semiconductor cost not including the power supply is about $50 USD. The parts are also available from MCM Electronics, Farnell and Newark Electronics. Since they are all the same company, these parts can be purchased just about anywhere in the world.

There are no recommended substitutes. No American manufacturer makes 900V PNP or NPN transistors with a low Cob anymore. Neither does Phillips of the Netherlands. The only manufacturers of these transistors are Sanyo and Toshiba, and only because they are heavily used in dynamic focus applications for large CRT monitors.

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The enclosure is a Mod.U.Line by Precision Fabrication Technologies Inc. (part number 03-1209-BW) and is available from Newark Electronics, probably Allied too. It measures 3″ x 12″ x 9″.

The Results

I do not think that the Omega II headphones can be damaged by this amp unless the bias is set way too high. If the bias is set right, the outputs are close to 0V at idle, and all the LEDs are lit, then the amp pretty much has to be working correctly. Now if one or more of the outputs is stuck at +300V or -300V, then something is seriously wrong and needs to be fixed. An oscilloscope really helps.

The amp can output 800Vp-p or 1200Vp-p with headroom. At 800Vp-p, THD is less than .008% from 20Hz to 20kHz. The actual frequency response is 0 to 45khz (-3db at 45kHz) into an Omega II load. Compared to the sound of my previous tube amplifier, the bass is no longer tubby; it’s very sharp and tight. The high end is no longer rolled off, so female voices sound much more real. If the bias supply is reduced to 280V, the amplifier will drive all electrostatic headphones. I tried it last night on a pair of SRX’s. I never ever heard them sound so good.

Last weekend, I took home a standard dummy head, and measured the SPL in Omega 2 headphones driven by this amplifier. With a drive signal of 800 volts peak to peak per side, the resulting spl is 106db. THAT’S LOUD! The amp can put out 1200 volts peak-to-peak, and thats louder! I just ordered a pair of Stax SR-001 MkIIs, which can reach up to 120dB. My ears distort before the amplifer/headphones do. It is quite loud at clipping, but the clipping is a hard clip with no oscillation or ringing. To use the amplifier with electret headphones, delete the bias voltage. And probably keep the output swing under 200V. Electrets phones when driven with this amplifier can probably get very very loud.

[Editor: Contact the author to discuss the possibility of obtaining pre-etched PC boards for this amplifier.]

c. 2000, Kevin Gilmore.
From The Homepage of Kevin Gilmore. Republished with permission.

Addendum

2/21/01: Corrected mislabeled transistor part number: 2SC1815 (was 2SA1815).

9/5/01: Corrected mislabeled transistor part number: 2SC3675 (was 2SC367).

2/12/2002Richard Albers built the following version of the CDEA amp with some interesting modifications of the original circuit. He writes:

I have changed the 2SK389 FET for a MAT02 Dual Transistor in the first Stage of the CDEH-Amp. There were no problems, and it all worked fine from the start. It sounds much cleaner then with the Dual-Fets now, and I guess they add less harmonics to the music.

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In the third Stage of the CDEH-Amp, I have changed the Voltage-Divider 350K/20K, which sets the Bases of the SC3675 at ca. 20V. For the 20K Resistor i have put in a 20V, 1.3W zener. For proper working, I set the current through the zener at 7mA. Two 25K ohm, 5W Mills non-inductive wirewounds replace the 350K, dissipating ca. 2.2W of heat. To reduce the zener noise, I have put a 4.7uF tantalum together with a 47uF electrolytic capacitor in parallel with the zener diode. Noise is no problem.

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The Cabinet is a very simple construction, with the advantage of ease changing components or parts. There is only a wooden base with two side-panels. The front is the large heatsink together with an aluminium-angle. A suitable top-cover is under construction. The whole construction could be made way smaller, all parts on one pcb, with a smaller toroid-transformer, and all built in a industrial case, but for my own usage, it’s ok.

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The two smaller transformers under the wooden cover are the 10H-chokes for the high voltage power supply. The little transformer on the bottom generates the bias-voltage. The oversized big-one is a special-made 250W transformer, from Experience-Electronics in germany. The electrolytics are from EPCOS (Siemens).

This is a further way to tune-up this fantastic machine. Together with the MAT02 dual-bipolar input device and using only the best parts you can get, such as non-inductive Caddocks, very low ESR electrolytic caps in the high voltage section, and so on, there is no better electrostatic headphone amp in the world. It sounds just fantastic!