A Simple Tube/Opamp Hybrid Amplifier.

by Alex Cavalli, Mark Lovell and Bill Pasculle

cavalli2_ampshot.jpg

Introduction

After seeing many of the excellent and eye-catching tube-solid state amplifiers of others, we’d like to present a slightly different topology of hybrid amplifier design, using the same two basic components of a tube and an opamp. This amp is simple and has less than $50.00USD worth of parts and no lethally high voltages. It makes an ideal tube headphone amp for those who like the sound of tubes but worry about the safety of high voltage equipment.

Bill was looking for an amp that sounded good and also was a good companion for his Rio Carbon portable player. He also suspected that others might like something similar as well. With high bit-rate MP3 or WMA files, the Carbon can produce excellent sound, but as with many other portable players, benefits by the addition of a decent headphone amp. He also wanted to use the amp in his cubicle at work – requiring good sound without taking up too much space.

The design of the amp started when Bill suggested that he wanted to build one of the YAHA (Yet Another Hybrid Amplifier) amps , but using a different design. Alex, being mostly interested in tube amps, was not initially interested but as Bill and Mark began to suggest requirements for the amp and some options for the input tube, it was obvious that there was room to improve the typical hybrid topology. After more thought, Mark provided a list of some requirements for the design that he felt were necessary to meet in order to improve on the hybrid topology.

Alex went to work taking the design discussions and turning them into a draft design. Our first name for the amp (as a joke) among the team was Stoopid Opamp Headphone Amp (SOHA). The name, as so often happens with skunk-works project names, eventually stuck and finally we are just calling this amplifier the Stoopid or the SOHA. Another name for the amp, with the same acronym, might be the Simple Opamp Hybrid Amplifier.

The original prototypes assured us that it is possible to make a surprisingly good performing amp utilizing a tube at relatively low voltage and while still keeping the build cheap, easy, and reasonably electric shock-free. After altering some of the power supply and circuit values and testing the prototype, we ended up with something that was stable and fun to listen to for extended periods. It applies some compression but that’s part of its charm, and it would be a rather sterile sounding amp without the 12AU7/ECC82 altering the sound the way it does with its 40V plate voltage.

Amplifier Circuit

Most of the hybrid amps that have appeared in HeadWize threads and elsewhere (such as the Millet hybrid) have used the same B+ for both the tube and the opamp. Some of these amps are designed to be portable enough to run from a battery, but most are really constrained to a low voltage DC supply of some kind plugged into the line and so are not truly portable.

In addition, it is generally true that tubes that are not designed for low voltage use will not perform well at 12-24V (which is why the Millet uses special low voltage tubes), so we decided to try to provide the tube with higher B+ to get better performance, while still keeping the voltages fairly low. This meant that the amp could be small and portable although requiring AC power. Like the other hybrid amps, the SOHA is designed to give the sound of tubes while avoiding the high voltage risk that some builders don’t like. Still, providing a higher B+ permits us to get excellent sound from a more commonly available tube like the 12AU7/ECC82, which is in good supply from NOS and current production sources and which gives a wide variety of choices for tube rolling. Having this wide selection also makes it easier for the amp to be constructed in any part of the world.

The way in which most other hybrids use a common B+ for both tube and opamp has two detrimental effects on a hybrid amp:

  1. It puts the opamp in a single ended configuration where it needs an output cap to block half the B+
  2. It forces the B+ on the tube to be low so as not to exceed the maximum opamp rail voltages.

The first design decision was to decouple the B+ for the tube and opamp. Doing this makes it possible to use a standard bipolar supply for the opamp, eliminating the large output cap (as in the Chu Moy’s pocket amp for example) and requiring only a small coupling cap between stages (see power supply discussion below).

The tube is loaded with a constant current source (CCS) for two reasons:

  1. The resulting non-linear performance associated with low voltage operation is partially offset by the high dynamic impedance of the CCS
  2. A CCS has a much better PSRR than a simple resistor making it possible to have more ripple in the B+ and, therefore, simplifying the B+ PS.

The original amp is designed to run with FET input opamps. See note 1 (the BJT opamp section) for a version using BJT input opamps.

After extensive prototyping by Mark and Bill, and after posting the first design to the HeadWize forums and receiving feedback from several builders, we modified the original design. The most notable change was to the heater circuit. Originally the heater voltage was supplied directly from the AC secondary with voltage dropping resistors. This approach was implemented initially to maintain simplicity. However, because there is so much variation in transformer regulation, line voltage, and heater characteristics, the simple resistors were replaced with a regulator circuit. A side-benefit is the elimination of power wasted as heat since the dropping resistors reached 105°C – 115°C under normal operating conditions.

The Amplifier Circuit

The basic amplifier circuit uses a very small number of components, as shown:

cavalli2_fig1.png

R1 100k Log Pot C1 1000u 10V Electrolytic
R2 300R 1/8W C2 100n 100V
R3 560R 1/8W D1, D2 1N4148 or similar
R4 2k Trimpot U1 OPA2134 or similar
R5 1M 1/8W V1 12AU7/ECC82 or equivalent
R6 150R 1/8W

Figure 1 – Basic SOHA Amplifier

The topology of the amp is a simple grounded cathode gain stage coupled through a capacitor to an opamp wired in unity gain mode. The standard SOHA uses LND150 depletion mode MOSFETs for the CCS for reasons discussed below. The amplifier circuit is, thus, very simple. Trim pots are provided as part of the cathode bias resistors to adjust for variations in tubes to set the plate voltages to ~40V. Each CCS is set to regulate at approximately 1mA.

An advantage of this design over many of the other hybrid designs is that there is no large coupling electrolytic at the output. The required inter-stage coupling capacitor is small making it possible to use good quality film/audio capacitors here at nominal additional cost.

With a 12AU7/ECC82 the input stage has a gain of about 12. This is sufficient for almost any source driving almost any headphones which is why the opamp is simply operating as a unity gain current buffer. However, with such high gain it is possible to exceed the input voltage tolerances for the opamp with just 1Vp at the input. The data sheet for the OPA2134 (and many similar opamps) indicates that the maximum input voltage is (V-) – 0.7V to (V+) + 0.7V. This means that the input swing must not exceed the supply voltage by more than one diode drop. The diodes ensure that this does not happen.

If the diodes conduct, the excess current passes into or out of the bipolar power supply. What happens thereafter depends on the ability of the tube to source/sink current and the ability of the PS to sink/source it. In this case, the tube will source/sink in the range of hundreds of micro amps which will find their way to the output caps of the bipolar supply which are in turn supplying current to the opamp V+ and V-. The output of the regulators will fluctuate some, but at this point the amp would not be operating properly anyway.

The standard CCS for the basic SOHA uses a single LND150 MOSFET in this configuration:

cavalli2_fig2.png

R7 1k 1/8W R8 360R 1/8W

Figure 2 – Standard LND150 MOSFET CCS

For a discussion of why this was selected as the standard CCS, see note 2 (the CCS comparison section) at the end of the article. One limitation on the standard SOHA CCS is the uneven availability of the LND150 MOSFETs globally. To ensure that this amp can be built almost anywhere, we have created two variations that use other devices for the CCSs. The first uses J113 JFETs and the second uses 1N5297 current regulator (CR) diodes. Here are the schematics for both variations:

cavalli2_fig3.png

R8

1k5 1/8W

Figure 3 – Alternate CCSs using JFETs and CRDs

Care should be exercised when building the SOHA with the J113 JFETs. Their maximum Vdss is 35V. Under normal operating conditions they will see only about 15-20V, but if the plate voltage on the tube is too low it is possible to exceed this maximum and destroy them. To protect the JFETS, the minimum plate voltage should never be set below 20V (see below the warning about adjusting the trimpots). The 1N5297 CRD has a 100V maximum and should withstand all of the normal voltages in this amp. The J505, noted in parenthesis, will also work but has only a 50V maximum. This makes the J505 a little more robust in this circuit than the J113, but less desirable than the 1N5297.

Power Supply Circuit

The key to this amp is the power supply. Initially the amp used an easy-to-acquire 30VCT/200mA transformer. As noted above, during the development and testing process, including builds by several HeadWizers, we changed the heater supply from AC to regulated DC. To accommodate the 150mA DC drawn by the heater it is necessary to increase the current spec on the secondary to 400mA. This will also give some headroom for the amp itself. Eventually, we chose the Amveco TE70053 toroid to replace the original split bobbin transformer. Another benefit to using the toroid is less EM radiation in the box and, since the SOHA also designed to be small, this reduces or eliminates problems with PS buzz. Other transformer possibilities are in the Power Supply section below. You can use a higher current rating transformer without difficulty, but if you increase the voltage be careful about not exceeding the maximum input voltage for the regulators. The bipolar opamp supply is a conventional regulated supply using 78L12/79L12 inexpensive regulators. They have a maximum input voltage of 40V.

The trick to the power supply is the use of a 1.5x full-wave voltage multiplier to generate the B+ for the tube. To make the voltage multiplier, the entire secondary of the transformer is rectified through a pair of coupling capacitors and bootstrapped on top of the V+ of the bipolar supply. With a typical transformer with 25% regulation and with no load on the B+ for the tube, this generates over 80V (this is marginally dangerous and will give you a pretty good sting so be careful).

The power supply, including the heater circuit is shown below:

cavalli2_fig4.png

R9 2k2 1/8W BR1, BR2 100V 1A Bridge Rectifier
R10 1k3 1/8W VR1 12V Fixed Regulator 78L12
R11 11k 1/8W VR2 -12V Fixed Regulator 79L12
C3-C6 100u 100V VR3 Adj. Negative Regulator LM337
C7, C8, C11 47u 16V D3, D4 1N4002
C9, C10, C12 470u 35V T1 30VCT 400mA Transformer

Figure 4 – the SOHA Power Supply

When the B+ is loaded with the tube, with each triode drawing ~1mA, the voltage is pulled down to between +55-65V. This means that there is plenty of headroom in the B+ to run the tube at+40V while still leaving space for driving 7-10V into the opamp. And this seems to give very good performance. As noted above, using a CCS for the plate load relieves ripple requirements on the B+ so a much simplified and less expensive filter section becomes possible. For the components as drawn the B+ ripple is about 1mV. The capacitor values are kept low and, hence, the capacitors are small and inexpensive. Again, for CCS PSRR comparisons see below.

The heater supply uses a full-wave rectifier into a negative 12.6V regulated supply. Pay careful attention to the orientation of the rectifying diodes. The heater supply is attached to the negative half of the bipolar supply. This was done because the heater current will pull down the input to the filter section of whichever half of the bipolar supply to which it is attached. Since we are using the positive supply to bootstrap the B+ for the tube we don’t want the heater supply to pull this voltage down. Therefore, it is derived from the negative supply because if the negative input to filter drops by a volt or two the regulator will not be affected. Pay careful attention to the orientation of the rectifier diodes and capacitors in the heater circuit since it is a negative supply. A power indicator LED can be attached to the heater supply taking care to note the polarity. The negative regulator should be heatsunk to dissipate about 2W.

Construction

The full schematic for both channels and the PS is shown below with the complete parts list less some miscellaneous components such as enclosure, power switch, etc.

cavalli2_fig5.png
Click here to see full-size schematic.

R1 100k Log Pot C3-C6 100u 100V
R2, R12 300R 1/8W C7, C8, C11 47u 16V
R3, R13 560R 1/8W C9, C10, C12 470u 35V
R4, R14 2k Trimpot D1, D2, D5, D6 1N4148 or similar
R5, R15 1M 1/8W D3, D4 1N4002
R6, R16 150R 1/8W U1, U2 OPA2134 or similar, dual or single
R9 2k2 1/8W V1 12AU7/ECC82 or equivalent
R10 1k3 1/8W BR1, BR2 100V 1A Bridge Rectifier
R11 11k 1/8W VR1 12V Fixed Regulator 78L12
R7, R17 1k 1/8W VR2 -12V Fixed Regulator 79L12
R8, R18 360R 1/8W VR3 Adj. Negative Regulator LM337
C1, C13 1000u 10V T1 30VCT 400mA Transformer
C2, C14 100n 100V

Figure 5 – Full SOHA amplifier, both channels and PS

For the J113 version eliminate R7, R17 and change R8, R18 to 1k5 1/8W. For the 1N5297 version simply replace the entire CCS with the single diode.

The amp has been built several ways by different HeadWizers [click here to see forum member Neurotica’s (Jim Eshleman) SOHA build narrative]. Mark and Bill initially built the prototypes using point to point wiring on perfboards and several others did so as well. Bill eventually also made a homemade PCB while Alex designed a PCB using the commercial package ExpressPCB (see below). Most builds to date have been like the prototypes with the PSU and amplifier circuits on the same board. Pictured below is a pictorial drawing showing how the SOHA can be wired point-to-point on a 4 x 6-inch perfboard.

cavalli2_fig6.png
Figure 6a – Bill’s SOHA constructed by point to point wiring on a perf board
Click here to see full-size layout.

Part of our purpose with the design and component specs is to keep everything as small and cheap as possible. The parts list shows the parts from the usual American suppliers. Mark was able to source similar parts from Farnell and RS in the UK.

Part # Mouser Catalog Number Description Qty. Price Total
R9 270-2.2K-RC Xicon 2.2k 1/8W

10

0.11

1.10

R3, R13 270-560 Xicon 560R 1/8W

10

0.11

1.10

R11 270-11K Xicon 11k 1/8W

10

0.11

1.10

R10 270-1.3K-RC Xicon 1.3k 1/8W

10

0.11

1.10

R5, R15 270-1.0M-RC (regular CCS) Xicon 1.0Meg 1/8W

10

0.11

1.10

270-100K-RC (mu follower) Xicon 100k 1/8W

10

0.11

1.10

R2, R12 270-300 Xicon 300R/1/8W

10

0.11

1.10

R6, R16 270-150-RC Xicon 150R 1/8W

10

0.11

1.10

R4, R14 652-3306K-1-202 Bourns 6mm 2K pot

2

0.56

1.12

C7, C8, C11 140-HTRL16V47 Xixcon 47uF/16V

3

0.07

0.21

C3, C4, C5, C6 140-HTRL100V100 Xicon 100uF/100V

4

0.42

1.68

C9, C10, C12 140-HTRL35V470 Xicon 470uF/35V

3

0.3

0.90

C1, C13 140-HTRL16V1000-TB Xicon 1000uF/16V

2

0.25

0.50

C2, C14 1429-1104 Xicon 0.1uF (100nF)

2

0.44

0.88

CCS Options
512-J113 J113 JFET

4

0.24

0.96

R8 270-1.5K Xicon 1.5k 1/8W

10

0.11

1.10

OR
689-LND150N3-G LND150 MOSFET

2

0.55

1.10

R8, R18 270-360 Xixon 360R 1/8W

10

0.11

1.10

R7, R17 270-1K-RC Xixon 1k 1/8W

10

0.11

1.10

  OR
610-1N5297 1N5297 CC Diode

2

4.29

8.58

BR1, BR2 821-DB102G 1A 100V Bridge

2

0.33

0.66

VR1 512-LM78L12ACZ LM78L12 Regulator

1

0.27

0.27

VR2 512-MC79L12ACP LM79L12 Regulator

1

0.34

0.34

VR3 512-LM337T LM337 Regulator

1

0.5

0.50

567-273-AB Wakefield Heatsink

1

0.38

0.38

Power LED 351-3310 Xicon Blue Led 3mm

1

1.5

1.50

271-560-RC 560R 1/4W LED Resistor

10

0.09

0.90

D1, D2, D5, D6 78-1N4148 1N4148 Diodes

10

0.03

0.30

D3, D4 512-1N4002 1N4002 Diodes

5

0.10

0.50

R1 313-1240-100K Taiwan Alpha 12mm pot, 100k

1

2.84

2.84

575-393308 IC Socket

1

0.36

0.36

J3 161-3502 3.5mm Headphone Jack

1

0.92

0.92

J1 161-1052 RCA Jack Black

1

0.82

0.82

J2 161-1053 RCA Jack Red

1

0.82

0.82

Total

38.04

OR
DigiKey
U1, U2 OPA2134PA-ND OPA2134PA

1

2.63

2.63

T1 TE70053-ND Amveco 30V CT 500mA

1

12

16.22

Total

18.87

Optional Sources
Newark
U1, U2 75C4624 OPA2134-PA

1

2.37

2.37

18C6948 J113 JFET

2

0.2

0.40

VR1 34C1091 LM78L12ACZ Positive regulator

1

0.28

0.28

VR3 34C1076 LM337T Regulator

1

0.67

0.67

R4, R14 46F1092 Bourns 6mm 2K pot

2

0.27

0.54

Antique Electronic Supply
V1 T-12AU7-JJ JJ 12AU7 Tube

1

8.95

8.95

V1 P-ST9-511 Tube Socket

1

1.95

1.95

Miscellaneous
Knob
Power Switch
Wire
Fuse holder and Fuse (0.25A)

The Amveco toroidal transformer (30VCT/500mA) is available from Digikey. Remember with 15-0-15 VAC (nominal) secondaries and the poor regulation of these inexpensive transformers, you will see over 21V with no load at the inputs to the bipolar power supplies and ~85V with no load for the B+. Because of the poor regulation make sure to use capacitors with voltage ratings that meet these off-load conditions. Note that the PS parts table shows 100V capacitors for the B+ section. If you use a higher voltage transformer make sure that the input to the regulators does not exceed their maximums (typically about 37V).

Some other possible split bobbin transformers are: Dagnall D3019 (0-240 pri), D3023 (0-115,0-115 pri). Both are 12VA. Other possible toroids are: MULTICOMP MCTA015/15 (0-115,0-115 pri), MULTICOMP MCFE015/15 (0-230V pri), or MULTICOMP MDCG015/15 (0-230V pri).

This design is optimized for 12AU7/ECC82 and its exact equivalents (5963, 6189, and 6680) rather than a close equivalent (or other types of tubes such as 6922).

All three flavors of CCS provide a degree of PSRR and some immunity from power fluctuations. They differ in availability worldwide and in maximum voltage ratings. The best overall CCS uses the LND150 MOSFET which is not available everywhere. The J113 FET is widely available but its maximum DC voltage is only 35 volts. Normally the FET wouldn’t see more than 15-20V unless the plate voltage gets too low. The 1N5297 CRD has a maximum voltage of 100 VDC but is not as widely available and is also expensive. Nevertheless, working amps have been built using all three types of CCS. Trim pots located at the cathodes are used to adjust the plate voltage. In order to prevent burning out the CCS FETs the cathode trim pots should always be turned to their highest resistance when swapping in a new tube.

OPA2134 and its relatives are fairly common opamps for audio. This was a good place to start. The authors would like to know how other FET input opamps perform and welcome feedback from builders. The OPA551, for example, is a FET input opamp that drops right into the Stoopid. However, it only comes in single packages so you will have to account for this with the build.

FET input opamps are preferred because there is a risk that BJT input opamps may tend to excessively load the tube and defeat the effect of the CCS. To use BJT input opamps see the section below for modifications to do this. The authors welcome feedback on the performance of the SOHA with BJT input opamps.

A BUF634 could easily be put into the unity gain feedback loop of the opamp to give super high output. One change that might be necessary if really trying to pull 200mA is to increase the size of the input capacitors in the bipolar PS to more like 2200uF. Even larger values may be required to get full bass.

Mark added 150 Ohm resistors (R6) at the outputs as this enables the amp to drive low and high impedance headphones without experiencing a large change in volume. These can be left out of the circuit at the builder’s discretion, however, their use is recommended. Likewise the pairs of 1N4148 diodes connected to the non-inverting inputs of the opamps are optional, but serve to protect the opamp inputs from overload and their use is recommended.

Here are a few details to pay attention to during construction and double check before applying power to your SOHA:

  • Wiring the Triad transformer is not intuitive; the pins are not numbered consecutively. Study the datasheet carefully.
  • The 78L12 and 79L12 do not share the same pinout.
  • The capacitors in the heater supply (as well as those in the negative half of the bipolar supply) have their positive leads grounded.
  • Use of a star-ground is highly recommended.
  • Use of shielded cable from the input jacks to the pot, from the pot to the tube grids, and from the opamp to the output jack is highly recommended. Attaching the safety ground to the star ground is optional. Most builds have worked fine with the star ground floating but an occasional unit has been quieter with the safety ground connected to the star ground.
  • Grounding the pot body is usually required to eliminate static/hum.

Wire dress is important in this amp to avoid hum. Keep all signal wires away from the transformer; keep the filament wires as far away from the audio circuit as possible.

PC Boards

We’ve created Express PCB boards for the SOHA. These are related to the full schematics with part numbers shown.

For the J113 version eliminate R7, R17 and change R8, R18 to 1k5 1/8W. For the 1N5297 version simply replace the entire CCS with the single diode.

ExpressPCB and PDF files for both the amp and power supply are included below. The tube socket on the amp board is in the center of the board. Note that the tube socket mounts on the foil side of the board. With this configuration you can easily mark a hole in the center of the standoffs and punch it out to pass the tube through so that the tube can stick up through the chassis while the components are sticking downward.

The copper layer in these PDF and ExpressPCB files can be used for home etched boards.

SOHA Amplifier Board (PDF)
SOHA Power Supply Board (PDF)
SOHA Amp and PS boards (ExpressPCB)

Techniques for making home PCBs were suggested by HeadWizer Bill Blair. Here are some links that Bill used to make his own SOHA boards using the single layer PDFs:

EasyPCB Fabrication
HomeBrew Printed Circuit Boards

The boards can be jumpered to use all three versions of the CCSs and to operate as standard plate drive or as source follower drive. This is the stuffing guide for these possible configurations.

cavalli2_fig12
Click here to see full-size stuffing guide.

Figure 6b – Bill’s stuffing guide for the SOHA amplifier PCB

Setup

Wire everything up but don’t put the tube/opamp in yet. Measure the voltages at the B+, V+, V-, and heater. They should be >80V, +12V, -12V, and -12.6V respectively. If they are not then there is a problem that must be fixed before inserting either the tube or the opamp.

If voltages are good and nothing has fried, power down and then insert the tube and opamp. Before powering up again, dial your trim pots so that they are in the maximum resistance position. This will put the maximum bias on the tube. Measure the voltage at the plates (pins 1 & 6) and adjust the associated trim pot until the voltage comes down to 40V for each plate. After these adjustments, measure the B+ again. It should be between 55-65V. Occasionally you may find a NOS tube does not work well in this circuit. You may need to replace the tube to get good results. If so, the tube is probably outside of its published operating characteristics. Each triode of 12AU7/ECC82 draws only 1mA from the B+ supply and at these low currents there can be a wide variation in operating characteristics, particularly among tubes that may be marginally within spec.

Results

OK, how does it sound? Well, in short, stoopidly good. When first powered up, the prototype plate voltage was only 17V and the amp sounded decidedly solid state. Very “steely” and just tonally “off”. As the plate voltage was raised the sound became more lush and tube-like. At 40V the amp began to perform extremely well. The SOHA easily rivals the Cavalli-Jones/Morgan Jones which costs over three times more to build! It’s got decent amounts of bass, classic sweet tube midrange and plenty of top end extension. Also the soundstage is extremely wide and respectably deep. This thing is just plain stoopid fun to listen to!

The amp drives headphones of any impedance between 16 Ohms and 300 Ohms without problems, which covers most that are currently available.

The compression applied by running the 12AU7 with 40V at the plate allows an unexpectedly refined sound with no sharp edges, yet without being mellow. It will reproduce transients when required and has a respectable dynamic range. The overall result is something than can be listened to for extended periods with no “listening fatigue” and providing a pleasingly wide and reasonably deep sound stage.

As is the case with tube amps, a warm up time is required and in this respect the authors agree 20 minutes is required for it to sound its absolute best, but of course it’s up and running after 30 seconds.

Mark has compared this amp to three other headphone amplifier designs available at HeadWize having built them: namely the CJ, the CL MkII, and the BCJ MkI, (the BCJ MkII was unavailable). All of the alternative designs used for comparison tests are more expensive to build, all require potentially lethal voltages, and all are optimized to ensure the tubes are working at their optimum.

Clearly, the SOHA would be the worst of the bunch? Not so. It proved itself to equal the CJ and gets closer than expected to the CL MkII. That’s pretty impressive stuff for a tube amp deliberately designed to be cheap and not use lethal voltages.

Tube rolling in this amp is also a lot of fun. Mark and Bill, who listen mostly through Sennheiser HD-600’s, found that grey-plate 5963’s from GE and RCA and Brimar sounded better than other tubes. Some other Headwizers with low-Z cans seemed to prefer black-plate versions of these tubes. Among the new production tubes, the Electro-Harmonix 12AU7 seemed to approach (but not exceed) the performance of the NOS tubes while the JJ 12AU7 was a somewhat distant second. Differences between tubes seemed to be in the clarity of the top end and the amount and quality of the bass.

Here are some photos of Bill’s SOHA in its final home.

cavalli2_fig7.jpg
Figure 7 – Bill’s SOHA Top Side and Figure 34 – Bill’s SOHA The Guts

Note 1: BJT-Input Opamps

As noted above the SOHA was designed to use FET input opamps. However, to permit opamp rolling, we’ve created two minor variations that permit the use of BJT input opamps.

Bipolar-input opamps like the TSH22IN, NE5532, or NE5534 can substitute for the 2134. But bipolar opamps will have lower input impedance than the FET input opamps. This will increase the loading on the tube and increase the distortion.

One way around the increased loading is to configure the CCS as an active load source follower. This variation requires only one change in wiring at the CCS and will only work for the LND150 and the J113 versions. An active load source follower is a variant of a well-known tube topology where the CCS that is acting as a plate load is also utilized as the output device in a follower configuration. With this topology the output impedance of the gain stage drops considerably and its ability to supply current increases in proportion. With both FET topologies we can wire the FETs as source followers to make a hybrid follower configuration for the first stage.

If you’re using the LND150 CCS you can convert the CCS into a source follower by simply changing the point where the coupling capacitor is connected. The FET then becomes a source follower with low output impedance and the ability to drive higher currents into the load.

cavalli2_fig8.png
Figure 8 – Changing the LND150 CCS for BJT-input Opamps

If you’re using the J113 CCS you can covert it to a source follower using the same technique:

cavalli2_fig9.png
Figure 9 – Changing the J113 CCS for BJT-input Opamps

Although the 1N5297 CRD is actually a JFET wired as a CCS we cannot access the source of the device so the CRD cannot be used when driving BJT opamps.

For example, a full amplifier schematic for the standard LND150 CCS with BJT opamp is:

cavalli2_fig10.png
Figure 10 – Driving BJT input opamps

If your amplifier exhibits high DC offset with BJT opamps, you can decrease the value of R5 from 1M to 100k or even 50k without overloading the gain stage. Note that decreasing the value of R5 while leaving C2 at 100nF also reduces the low frequency response of the amplifier. To correct for this, increase the value of C2 so that the product of R5 x C2 is the same as 1M x 100nF. For example, if you decrease R5 to 100k, then to maintain the same low frequency response, increase C2 to 1uF.

For BJT input opamps, the full schematic is this:

cavalli2_fig11.png
Click here to see full-size schematic.

R1 100k Log Pot C3-C6 100u 100V
R2, R12 300R 1/8W C7, C8, C11 47u 16V
R3, R13 560R 1/8W C9, C10, C12 470u 35V
R4, R14 2k Trimpot D1, D2, D5, D6 1N4148 or similar
R5, R15 100k 1/8W D3, D4 1N4002
R6, R16 150R 1/8W U1, U2 BJT opamp, dual or single
R9 2k2 1/8W V1 12AU7/ECC82 or equivalent
R10 1k3 1/8W BR1, BR2 100V 1A Bridge Rectifier
R11 11k 1/8W VR1 12V Fixed Regulator 78L12
R7, R17 1k 1/8W VR2 -12V Fixed Regulator 79L12
R8, R18 360R 1/8W VR3 Adj. Negative Regulator LM337
C1, C13 1000u 10V T1 30VCT 400mA Transformer
C2, C14 1u 100V

Figure 11 – Full SOHA with BJT opamp, both channels and PS

Note the part changes shown in red. These are the only changes necessary to use BJT opamps in the SOHA. The layout of amp does not change.

Note 2: CCS Comparisons

The choices for standard CCS and acceptable variations are derived from three criteria:

  1. maximum breakdown voltage of the CCS
  2. current regulating ability
  3. PSRR

These comparisons were done using PSpice simulations. These simulations are not likely to give absolute accuracy, but they are good at providing a relative comparison among the various CCSs.

Simulations were done for the following CCS types:

  • Single LND150
  • Single J113
  • Single PN2907A
  • Single 1N5297
  • Single PN2907A with CRD bias string
  • Cascoded J113
  • Cascoded PN2907A
  • Cascoded PN2907A with CRD bias string

This table shows the current variation, PSRR, and breakdown voltage (BV) for these various configurations:

Current Variation

Ripple at Plates

BV

300mVp 1kHz Input

1mVp 120Hz Ripple from PS

Delta I (uV)

Delta V (mV)

PSRR (db)

Topology

Cascoded JFETs (J113)

0.01

0.002

-54

35

Single MOSFET (LND150)

0.9

0.0025

-52

500

Cascoded BJTs with CRD

0.26

0.0095

-40

60

Single BJT w/ CRD

1.2

0.0135

-37

60

CRD (1N5297)

6.6

0.017

-35

100

Single JFET (J113)

13.3

0.034

-29

35

Cascoded BJTS (PN2907A)

0.34

0.058

-25

60

Single BJT (PN2907A)

1.2

0.06

-24

60

The cascoded JFETs have the best current regulation, followed by the MOSFET. The cascoded BJTs with CRD and without CRD have the next best current regulation. This might make these the next best choices. But, we must look at the PSRR and BV tables too.

The cascoded JFETs also have the best PSRR but they have a low BV. The LND150 has nearly the same PSRR (indistinguishable as a simulation result) but a very high BV. The LND150’s current variation comes in fourth behind the cascoded BJTs. However, the PSRR for the cascoded BJTs is 12db and 27db less than the LND150. Furthermore the BV for the BJTs is on the margin of where the voltages in the amp may be, and the BJT CCSs require many more parts than the either the JFETs or the MOSFET.

Taking all of these results together, the LND150 rises to the top for the standard SOHA because of its good current regulation, excellent PSRR, very high BV, and low parts count. The cascoded JFETs come in second because of their excellent regulation, PSRR, and low parts count. The CRD comes in third because of its good regulation, high BV and extreme simplicity (only one part).

Appendix: Simulating the Amplifier in OrCAD PSpice

Alex Cavalli has provided the project files for simulating this amplifier using OrCAD Lite circuit simulation software. The simulations will run in OrCAD Lite 9.1 or 9.2 only (later versions of OrCAD Lite and OrCAD Demo are more restrictive and will not run the simulations). The installation files for OrCAD Lite 9.1 or 9.2 can be downloaded from various educational sites on the internet. Search for them using the keywords OrCAD or Pspice and 9.1 or 9.2. OrCAD 9.1 is the smaller download (27MB). If you have trouble finding these files, email a HeadWize administrator for help.

There are 4 programs in OrCAD Lite suite: Capture, Capture CIS, PSpice and Layout. The minimum installation to run the amplifier simulations is Capture (the schematic drawing program) and PSpice (the circuit simulation program).

Download Simulation Files for SOHA Headphone Amplifier

After downloading cavalli2_soha_sim.zip, create a project directory and unzip the contents of the cavalli2_soha_sim.zip archive into that directory. Move the .lib and .olb files into the \OrcadLite\Capture\Library\PSpice directory. These are the component libraries containing the SPICE models for the vacuum tubes, MOSFETs and opamps used in the SOHA. (Note: heater connections are not required for any of the triode models.) In OrCAD’s Capture program, open the stoopid.opj project file.

The two basic types of simulation included are frequency response (AC sweep) and time domain. The time domain analysis shows the shape of the output waveform and can be used to determine the amplifier’s harmonic distortion. They both run from the same schematic, but the input sources are different. For the frequency response simulation, the audio input is a VAC (AC voltage source). The time domain simulation requires a VSIN (sine wave generator) input. Before running a simulation, make sure that the correct AC source is connected to the amp’s input on the schematic.

cavalli2_sim.png

The following instructions for using the simulation files are not a complete tutorial for OrCAD. The OrCAD HELP files and online manuals include tutorials for those who want to learn more about OrCAD.

Frequency Response (AC Sweep) Analysis

  1. Run OrCAD Capture and open the project file stoopid.opj, if not already open.
  2. In the Project Manager window, expand the “PSPICE Resources|Simulation Profiles” folder. Right click on “Schematic1-ac” and select “Make Active.”
  3. In the Project Manager window, expand the “Design Resources|.\cavalli.dsn|SCHEMATIC1” folder and double click on “PAGE1”.
  4. On the schematic, make sure that the input of the amp is connected to the V4 AC voltage source. If it is connected to V3, drag the connection to V4.
  5. To add the triode library to the Capture: click the Place Part toolbar button (orcad1.gif). The Place Part dialog appears. Click the Add Library button. Navigate to the triode.olb file and click Open. Make sure that the analog.olb and source.olb libraries are also listed in the dialog. Click the Cancel button to close the Place Part dialog.
  6. From the menu, select PSpice|Edit Simulation Profile. The Simulation Settings dialog appears. The settings should be as follows:
      • Analysis Type: AC Sweep/Noise
      AC Sweep Type: Logarithmic (Decade), Start Freq = 10, End Freq = 300K, Points/Decade = 100
  7. To add the triode library to PSpice: Click the “Libraries” tab. Click the Browse button and navigate to the the triode.lib file. Click the Add To Design button. If the nom.lib file is not already listed in the dialog list, add it now. Then close the Simulation Settings dialog.
  8. To display the input and output frequency responses on a single graph, voltage probes must be placed on the input and output points of the schematic. Click the Voltage/Level Marker (orcad2.gif) on the toolbar and place a marker at grid of U6. Place another marker above R9 at the amp’s output.
  9. To run the frequency response simulation, click the Run PSpice button on the toolbar (orcad3.gif). When the simulation finishes, the PSpice graphing window appears. The input and output curves should be in different colors with a key at the bottom of the graph.
  10. The PSpice simulation has computed the bias voltages and currents in the circuit. To see the bias voltages displayed on the schematic, press the Enable Bias Voltage Display toolbar button (orcad5.gif). To see the bias currents displayed on the schematic, press the Enable Bias Current Display toolbar button (orcad6.gif).

Time Domain (Transient) Analysis

  1. On the Capture schematic, make sure that the input of the amp is connected to the V4 sinewave source (VAMPL=0.4, Freq. = 1K, VOFF = 0). If it is connected to V3, drag the connection to V4.
  2. In the Project Manager window, expand the “PSPICE Resources|Simulation Profiles” folder. Right click on “Schematic1-signal” and select “Make Active”
  3. From the menu, select PSpice|Edit Simulation Profile. The Simulation Settings dialog appears. The settings should be as follows:
      • Analysis Type: Time Domain(Transient)
      Transient Options: Run to time = 80ms, Start saving data after = 40ms, Max. step size = 0.001ms
  4. To display the input and output waveforms on a single graph, voltage probes must be placed on the input and output points of the schematic. Click the Voltage/Level Marker (orcad2) on the toolbar and place a marker at grid of U6. Place another marker above R9 at the amp’s output.
  5. To run the time domain simulation, click the Run PSpice button on the toolbar (orcad3). When the simulation finishes, the PSpice graphing window appears. The input and output curves should be in different colors with a key at the bottom of the graph.
  6. To determine the harmonic distortion at 1KHz (the sine wave frequency), harmonics in the output waveform must be separated out through a Fourier Transform. In the PSpice window, press the FFT toolbar button (orcad7.gif). The PSpice graph changes to show the harmonics for the input and output waveforms. The input and output curves should be in different colors with a key at the bottom of the graph.
  7. The fundamental frequency at 1KHz will have the largest spike. The other harmonics are too small to be seen at the default magnification. In the PSpice window, press the Zoom Area toolbar button (orcad8.gif) and drag a small rectangle in the lower left corner of the FFT graph. The graph now displays a magnified view of the selected area. Continue zooming in until the harmonic spikes at 2KHz, 3KHz, etc. are visible.
  8. Harmonic spikes should exist for the output waveform only. The input is an ideal sine wave generator and has no distortion. To calculate total harmonic distortion, add up the spike values (voltages) at frequencies above 1KHz and divide by the voltage at 1KHz (the fundamental).

Note: simulations only approximate the performance of a circuit. The actual performance may vary considerably from the simulation as determined by a number of factors, including the accuracy of the component models, and layout and construction techniques.

c. 2006 Alex CavalliMark Lovell and Bill Pasculle (remove _nospam_).

A Pocket Headphone Amplifier.

by Chu Moy

cmoy2_1a

“Thank you for your amplifier design. I built it and can’t believe how wonderful it makes my AKG K340 headphones sound as well as my Sennheiser 600.”
– A DIYer.

While doing research for the article Designing an Opamp Headphone Amplifier, I built a portable headphone amplifier for testing purposes. Each channel uses a single Burr-Brown OPA134 opamp in a non-inverting configuration. It has adequate current capability to drive most headphones without an output stage. I have used it with Sennheiser 465s (94dB SPL) and achieved ear-splitting volume. The amplifier is ideal as a booster for power-conserving stereo sources such as portable CD players and for interfacing with passive EQ networks such as tone controls or a headphone acoustic simulator.

The Amplifier Design

cmoy2_2a
Figure 1

The schematic for one channel of the amplifier is shown in figure 1. All of the parts, except for the opamps, are available from Radio Shack. In several instances though, higher quality parts are available from other sources for about the same price that Radio Shack charges. The parts are commonly available, so look around for good buys. I do recommend Radio Shack’s 1/4W Metal Film Resistor Assortment (RS 271-309). It contains 50 resistors in popular values and nearly all of the values needed for this project. The total cost for this project should be no more than $20 – $25 US, assuming you already have general purpose items such as wire (I used solid 22 ga.).

The original opamp for this design, the OPA132, has been discontinued. The OPA134 is the audio-specific version of the OPA132 and will work identically in this circuit. It was selected for its excellent specs: FET inputs for high input impedance and low offset current, 8 MHz bandwidth, 20V/uS slew rate, ultra low noise, ultra low distortion, etc. It has fine PSRR (power supply rejection) numbers, can run on as little as ±2.5V (very important in a portable design) and includes built-in current limiting. The OPA134 costs less than $3.00 per unit from Digi-Key Electronics. It comes in a popular dual version: the OPA2134, which contains two opamps in a single package. Be sure to get the “DIP” package opamps; SOIC opamps are miniatures that are very difficult to handle.

Other opamps can be substituted, but make sure they will work with battery voltages (as little as ±3V) and are stable without external compensation. Also check the opamp’s current capability and current draw. The OPA134 has a quiescent current of about 4mA and will not drain the battery excessively. It can output almost 40mA into a short circuit at room temperature. Modern dynamic headphones need about 10mW to reach full volume. For more information, see Understanding Headphone Power Requirements.

cmoy2_17
Amplifier Frequency Response

The OPA134 is wired as a non-inverting amp with a gain of 11. At this gain, the output impedance of the amplifier is less than 0.2 ohms throughout the audio range. The high-pass filter C1-R2 at the input blocks DC current and has a corner frequency of about 15Hz. Substituting a 1uF capacitor will lower the corner frequency to 1.5Hz. However, 1uF capacitors tend to be too large for the recommended enclosure. Instead, if a lower corner frequency is mandatory, try increasing R2 to 1M (and scale R1 accordingly). You could omit C1 entirely, if DC input protection is not important. I recommend leaving C1 in the circuit.

If the amp will be driving low impedance headphones (32 ohms or less) such as the Grados, see appendix 1 for ways to optimize the amp for low impedance loads. R5 is an optional load resistor, which is explained in appendix 1. It can help reduce residual hiss and keep the power supply balanced.

The original pocket amp did not have a volume control, due to insufficient space in the enclosure (but see the next section for information on adding mini-pot volume control). Nor was a volume control necessary since the intended audio sources such as portable CD players and FM stereos already had volume controls. I did want the ability to reduce the input level as required to avoid overloading the amplifier (for example, some portable stereos have very high output voltage levels even when the volume control is set near 0). With R1 = 100K ohms, the LEVEL switch (SW1) drops the input voltage by 50% (6dB). At R1 = 470K ohms (the value I used), the switch attenuates the input by 15dB.

cmoy2_3
Figure 2

Several DIYers have written me to ask about adding a true volume control to the amplifier. In figure 2, R1 and SW1 are replaced with a dual, audio-taper mini potentiometer. The suggested pot values are 10K to 50K ohms. The enclosure in the prototype is barely 1″ tall, and the front panel is already crowded with and LED, switch and jacks. Mini dual pots are hard to find. Currently, Tangent’s Parts Shop is selling the ALPS RK097, a dual 10K audio mini pot, for a reasonable $3.25. Digikey sells the Panasonic EVJY10 series pots in 10K and 50K versions (part nos. P2G1103-ND for 10K, P2G1503-ND for 50K) for less than $3 each. The excellent dual 10K Clarostat 585 conductive plastic pots can be ordered from Newark Electronics (part no. 585DX4Q25F103ZP) for less than $3 each. Radio Shack sells a physically larger, dual 100K pot (RS 271-1732), which will work if the value of R2 is increased to between 200K and 1M. (C1 can remain at 0.1uF, and the threshold frequency of the high pass filter will decrease with larger values of R2.)

cmoy2_4

The diagram above shows how to wire the Clarostat and Panasonic pots. The ALPS pot has the same wiring as the Clarostat. Use an ohmmeter to confirm the wiring diagram. First, choose one section of a dual pot to check. Connect an ohmmeter to measure the pot resistance from the middle terminal (wiper) to one of the end terminals. Then monitor the meter as the pot shaft is turned clockwise from minimum to maximum. If the resistance increases as the pot shaft is turned clockwise, then the end terminal being measured goes to the amplifier ground. If the resistance decreases as the pot is turned clockwise, then the other end terminal should be grounded.

The Power Supply

cmoy2_5
Figure 3

The power supply circuit (figure 3) converts the 9V battery into a ±4.5V dual supply. Although the OPA134 could run from a single supply, it (and other opamps) are designed for dual supplies, and a dual supply is required for direct-coupling the output. This virtual ground sits at 4.5V, but works because opamps only care about relative power supply voltages. At idle, the opamp output is still 0V (minus a millivolt or two of offset) without capacitor coupling. However, if the headphone amp will also double as a preamp, add a capacitor to the opamp output to block DC, if the input stage of the power amplifier is direct coupled.

The left and right channels are connected in parallel to the power supply. Choose the largest filter caps (C1 and C2) that will fit in the enclosure. I used 220uF caps, but would gladly have replaced with 330uF or higher caps if my enclosure had been bigger. Appendix 3 below discusses power supply options in depth: adding dual 9V supply, making a battery pack, recharging 9V NiCad/NiMH batteries, choosing an AC adapter, etc.

Putting It Together

I assembled the circuit on a printed circuit, 3-hole pad protoboard. I used a Vector Circbord board from Mouser Electronics (Stock No. 574-3677-6). This Circbord has an excellent circuit pattern (featuring numerous bus strips throughout) for this project. Radio Shack sells non-solder-plated boards, which are an acceptable substitute, but the copper will oxidize in time. I cut a small square (about 2″ x 1.75″) of the protoboard with a utility knife to fit the case (mark a section on the board, score it several times with the utility knife and straight-edge, and then break off the section). When cutting the board, make sure to include at least 3 foil “buses” for the power supply and ground. I socketed the ICs using gold-plated machined-contact sockets which work with low insertion force.

cmoy2_13

The case is a PacTec HML-9VB (Mouser 616-62582-510-039 or 616-62578-510-000). It measures 2.75″ x 4.6″ x 1″ with a built-in 9V battery compartment and both opaque and transparent red plastic front panels. (Note: PacTec may discontinued the red panels). I chose the red plastic panel because it’s thinner and easier to mount the headphone jacks. Many DIYers have been using colorful candy mint tins as enclosures. If the tin’s interior is conductive, it must be insulated with electrical tape or it could cause short circuits. The headphone jacks are enclosed units for 1/8″ stereo plugs. Radio Shack sells a version of these jacks (RS 274-249). I ordered higher quality units that have spring-loaded contacts from Mouser Electronics (Stock No. 161-3502).

cmoy2_14a
Figure 4

The layout of the switches, jacks and the power LED on the front panel is shown in figure 4. The placements are a little tight, but I think it turned out well. By the way, the LED can be either a low current type or an ultra-bright type. It is biased at less than 1mA to conserve battery power and still produces a very strong light. I used a 5mm LED placed in a LED bezel (RS 276-079) before being mounted on the front panel.

cmoy2_15

Note: If the amplifier is housed in a plastic enclosure, the LEVEL switch must be grounded or the amplifier will hum when the switch is touched. To ground the switch, strip about 1.5″ of insulation from a 5″ length of 22 ga. solid wire, tin the exposed end if necessary, and tightly wrap the exposed end around the groove at the rear of the metal mounting flange of the switch, twisting the end to form a secure, closed loop. Trim the other end of the wire to a suitable length and solder it to the circuit ground. The same is true if a volume control replaces the level switch. If the pot has a metal shaft and the amplifier will be mounted in a plastic case, the pot housing may have to be grounded to prevent hum. Follow the same directions for grounding the level switch housing.

cmoy2_1

The project came together very quickly – about two evenings – and without incident. I attribute the quick assembly to the simple design of the circuit and the neat layout provided by the Vectorbord. The circuit was first built on a standard breadboard and then transferred to the Vectorbord. The amp worked immediately when the power was applied. I did tweak the power supply for improved stability. My amplifier does not have a belt clip, but add-on belt clips are available at Radio Shack.

The Results
cmoy1_15

The sound of the amplifier is excellent, with solid bass and a sizzle-free, detailed high end. It powered my Sennheiser 465 headphones effortlessly. A 9V alkaline battery can power the amp for several days of continuous play (high-capacity NiCad and NiMH rechargeable batteries will also work). When paired with my modified Linkwitz acoustic simulator, which is housed in an identical enclosure, the set make for a truly “dynamic duo”. I pack them and a CD player for travel in a Case Logic KSDM-1 case. Since the amp and acoustic simulator are lightweight, they are well-suited for people on the go who like to take with them a complete listening system (of course, you could build both projects into a single enclosure for even greater convenience). Given the low overall cost and the high quality parts used, this project “amply” rewards for the modest expenditure.

Appendix 1: Tweaking the Amp for Low Impedance Headphones

The OPA134 opamp produces a small DC offset voltage, which does not affect the amp’s performance when driving medium to high impedance headphones (over 100 ohms). Low impedance headphones (32 ohms or less) can cause the power supply to become unbalanced, because a small current flows though the load, even when the amp is at idle. This table compares the power supply voltages with the Sennheiser HD600 (300 ohms) and Sony MDR-G52LP (24 ohms) headphones connected to the amp.

Amplifier Load V+ V-
No headphone 3.9V -3.9V
HD600 (300 ohms) 3.9V -3.9V
MDR-G52LP (24 ohms) 4.2V -3.7V

Note: the battery by itself measured 8VDC.

There is disagreement about whether this almost negligible offset is worth the trouble to fix. With opamps other than the OPA134 series, the offset might be higher and the power supply imbalance could be greater. The offset has not damaged any of my headphones, but it might impact performance slightly by reducing the amp’s power output, injecting noise and/or draining the battery. To determine whether a certain headphone unbalances the power supply, measure the V+ and V- values with and without the headphones plugged in (and no music playing).

For those who want to reduce or block the offset current, figure A1 shows two ways to modify the amp for optimal performance with low impedance headphones: a) add a load resistor or b) AC-couple the amp’s output. A third way is to rebuild the power supply with an active virtual ground device like the TLE2426 or an opamp-based equivalent. Active virtual ground circuits are described in the addendum.

cmoy2_a1_2
Figure A1

Solution A is the simplest and allows the output to remain DC coupled. The load resistor (figure A1a) will help stabilize the virtual ground and reduce any hiss or noise in the system. The load resistor does create a voltage divider effect with low impedance headphones, and so may lower the amp’s gain and maximum output power and possibly alter the frequency response. Some say that the pocket amp’s gain of 11 is too high for low impedance headphones, so the small drop in gain due to R5 might be desirable anyway. Choose a R5 value just large enough to stabilize the power supply without too much volume loss. I recommend a 1/4 watt, metal film resistor in the 20-50 ohm range.

Solution B avoids a voltage divider effect because although the capacitor blocks DC current, it is largely invisible to audio frequencies. The circuit in figure A1b shows how to switch between AC-coupled and DC-coupled outputs for the highest fidelity with medium and high impedance headphones (the load resistor in solution A could be switched too). Choose the largest value electrolytic capacitor that will fit in the enclosure. A 220uF capacitor will give a flat response down to about 22Hz in 32-ohm headphones.

Use a high quality, low impedance electrolytic capacitor to minimize any sonic coloration. High quality electrolytic caps don’t have to be expensive. The Nichicon Muse KZ series 470uF, 25V sells for less than $1.00 at the time of this writing. The Panasonic FC and FM series caps are also less than $1.00 each. The exotic Elna Silmic II series (which feature a silk fiber dielectric instead of paper) has a 470uF, 25V unit for less than $2.00 each. By comparison, an ultra high-end type like the Black Gate 470uF, 16V typically sells for around $12.00 each and is not recommended for this amp.

Appendix 2: Ideas for Troubleshooting Noise

When built as recommended above, this amplifier is a quiet performer with virtually no background noise. It is more immune to EM and RF interference than some other amplifiers I have heard. The pocket amplifier remained quiet when tested near an old elevator facility that was known for generating loud crackles in another, more susceptible design. Nor did I hear any RF despite that the building had an internal RF communications system.

Nevertheless, there have been a few reports of problems with noise. The first step in troubleshooting noise is to make sure it is coming from the amplifier itself, and not from the audio source. Disconnect the audio source and listen to the pocket amp for any background hiss, static, RF (radio frequency) or EM (electromagnetic) interference. If the amp is driving low impedance headphones (32 ohms or less), try installing R5 (see figure 1) and/or AC coupling the amp’s output as described in appendix 1.

If the noise is primarily RF or EM interference and is not coming from the audio source, it is probably due to long interconnects and headphone cords, which can act as antennas that channel RF signals into the headphone amplifier. The easiest way to block RF noise is to place one or more clip-on ferrite noise suppressors on the audio cables. They should be located on the end of a cable as close as possible to the input or output of the headphone amplifier. The clip-ons can be removed if the interference is temporary and subsides. See A Quick Guide to Headphone Accessories for more information on ferrite clip-ons.

Another way to deal with RF/EMI interference is to shield the circuit either by putting the it in a steel or mu-metal enclosure (connect the circuit ground to the metal case) or by lining the interior of the plastic enclosure with a shielding foil (such as copper). The bottom of the case where the circuit board rests must be insulated with electrical tape to avoid shorting out the amp. If foil is used, it must be connected to the circuit ground. Copper foil shielding tape could also be used (stain glass supply retailers sell inexpensive copper tape).

DIYers have told me that the high gain of the pocket amplifier can emphasize hiss from noisy portable CD players or other audio sources, especially when driving low impedance, high efficiency headphones. If CD player hiss is a problem, try taking the CD output from the Line Out instead of the Headphone Out – in which case, the amplifier must be constructed with a true volume control instead of the LEVEL switch as discussed above.

cmoy2_a2.gif
Figure A2

Another option is to reduce the gain of the amplifier to minimize hiss. Try a gain between 2 and 6 (R3 = 10K ohms to 4.7K ohms). If the amplifier will also be used with higher impedance headphones that can benefit from higher gain, make the gain adjustable with a switch to select between different value feedback resistors (figure A2). Again, make sure to ground the metal housing of this feedback resistor switch to prevent hum and noise from the switch itself (see instructions for grounding the level switch above).

Appendix 3: Power Supply Options

cmoy2_6
Figure A3

There are several situations, where the pocket amp could benefit from a higher voltage power supply – when driving high impedance headphones, when the amplifier is being fed from a high gain equalizer or when the listener just wants more volume. With very high impedance headphones (600 ohms or more), the amp may not be able to develop sufficient voltage across the load for maximum power transfer. If the amp is fed from an equalizer or tone control with a high boost, the output of the pocket amp could be driven into clipping.

cmoy2_7a
Figure A4

In such cases, I recommend using a ±9V dual battery supply, which is nothing more than two 9V batteries in series (figure A3) or an external power source such as an AC adapter or battery pack (figure A4). R1 can remain 10K ohms, but any value between 10K and 15K ohms will work fine. Unfortunately, two 9V batteries will not fit in the specified enclosure for this project. The Pac-Tec model K-HML-ET-9VB measures 4.6″ x 2.75″ x 1.5″ and has a compartment for two 9V batteries (Newark Electronics part. no. 93F9946).

cmoy2_8
Figure A5

Figure A5 shows a simple 15VDC external battery pack consisting of 10 AA batteries in a battery holder. The battery holder is Caltronics model BH107 and has snap terminals which fit standard 9V battery snap clips. Radio Shack sells an 8 cell version (RS 270-387) which will output 12VDC. The cable can be any thin 2-conductor cable. I made my own cable by braiding 3 lengths of 24 ga. stranded hookup wire (2 black and 1 red). Only 1 red and 1 black wire carry voltage; the second black wire functions as a shield.

One end of the cable is terminated with a 9V battery clip (RS 270-324). The red wire from the battery clip will carry the (+) voltage when connected to the battery holder and is connected to the red wire of the cable. Only one of the black wires is connected to the (-) wire of the battery clip; the other black wire is not connected on this side. The other end of the cable is terminated with a submini (2.5mm) 2-conductor phone plug, such as the Switchcraft 850X (Mouser 502-850X). Wire the plug so that the tip carries the (+) voltage. The two black wires connect to the ground of the plug. Insulate any exposed connections with a thin layer of electrical tape.

cmoy2_9.jpg
Figure A6

The power jack is the matching submini (2.5mm) 2-conductor phone jack, closed circuit type, such as the Switchcraft TR2A (Mouser 502-TR-2A). The jack is wired so that when the plug is inserted, the internal 9V battery is automatically cut off (figure A5). If the 9V battery were not cut off, the higher external voltage would flow into the battery and possibly cause it to explode. Therefore, the wiring of this jack must be done very carefully. Use a voltmeter to test the jack:

With the jack unplugged and the 9V internal battery installed, the V+ output terminal should read about 9VDC.

Insert the plug (do not connect the battery holder) into the jack. The voltage at the V+ terminal should read 0V (meaning that the internal battery has been cut off).

Remove the internal 9V battery and connect the battery holder (with batteries) to the cable. The voltage at the V+ terminal should be about 15V (or 12V with the 8-cell holder). The voltage across the internal 9V battery clip should be 0V (meaning that there is no backflow of voltage into the battery).

cmoy2_10

The jack should be mounted in the upper right-hand corner at the rear of the enclosure’s cover. Enlarge the mounting hole of the jack, as necessary, so that mounting nut will be installed flush with the top of the insertion tube (see figure A5). Note: the mounting nut MUST be flush with the top of the jack’s insertion tube or the power plug will not seat properly – a dangerous situation that could short the battery pack. If either the internal 9V battery or external battery pack gets hot during use, there is short circuit somewhere. Disconnect the battery pack immediately and resolve the problem.

 

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The battery pack also could short if the plug were to come partially loose in the jack. For this reason, I do NOT recommend using this battery pack while traveling. Safer alternatives to the phono plug and jack are coaxial DC connectors, which will not short if the plug is unseated. When the amp was being constructed, I could not find DC coaxial jacks small enough to fit on the side of the case. The Switchcraft 712A (Mouser 502-712A, Jameco 281842) fits in a 0.313 inch hole. The mating plug must accept a 2.5mm (0.1″) pin, such as Switchcraft 760 (Mouser 502-760, Jameco 281877).

The mounting threads of the power jack are in electrical contact with the power jack’s ground. If the amplifer is put in a metal enclosure, the virtual ground and the power jack ground must NOT be connected together or the virtual ground will be shorted out. To prevent this occurrence, insulate the power jack’s mounting threads from the metal enclosure with nylon washers or electrical tape on both sides of and within the jack’s mounting hole. Use an ohmmeter to confirm that the power jack ground is not in electrical contact with the enclosure.

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An AC adapter could replace the external battery pack. Most AC adapters are poorly filtered and will introduce noise into the amplifier. The best AC adapter for this project is a wall-wart with a regulated, non-switching supply. The adapter shown above (RS 273-1662) can output up to 12VDC at 300mA regulated. It also comes with a set of interchangeable power plugs, including a 2.5mm phono plug that should be compatible with the power jack in figure A5, so long as the voltage polarity is correct.

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Figure A7

The circuits in figure A7 turn the AC adapter into a NiCad/NiMH trickle charger with a 20mA charging current. Trickle charging takes longer but is gentler on the battery. The circuits are identical except for the value of the resistor that sets the charging current. Figure A7a is for the specified NiCad battery, and A7b is for the specified NiMH battery. The 9V NiCad from Radio Shack (RS 23-448) has a capacity of 120mAh and should achieve a full charge (8.2V) in about 5 hours. The 9V NiMH (PowerEx MH-96V230 by Maha) has a higher voltage and almost double the capacity of the NiCad. It will take almost 10 hours to fully charge (9.6V). NiMH batteries are very sensitive to overcharging. The charger must be turned off when the battery is fully charged to avoid shortening the battery’s lifespan. The 1N4001 diode prevents the battery from discharging backwards if the 12V adapter is not turned on but is still plugged into the amp.

Appendix 4: Turning the Pocket Amp into a Personal Monitor

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Figure A8

Commercial personal monitors for musicians can be expensive, yet are essentially nothing more than headphone amplifiers with a limiter and/or a balanced input option. Figure A8 shows the pocket amplifier with both balanced and unbalanced inputs. This simple wiring trick for converting balanced signals to single-ended signals isn’t free: the signal amplitude is cut in half, but the loss can be compensated by turning up the volume. A true balanced converter that preserves the signal amplitude and noise rejection can be found in Designing an Opamp Headphone Amplifier.

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Figure A9

Figure A9 shows an adjustable clipper, which can limit headphone volumes to safe levels. The maximum voltage that the headphone can see is 0.7Vp (the forward bias voltage of the diodes), so the clipper is most effective with high efficiency headphones of low to medium impedance (less than 200 ohms). High impedance headphones may not achieve enough volume even at the maximum setting. In that case, try replacing each diode with two diodes in series to raise the clipping voltage to 1.4Vp. The clipping effect is a little harsh because of the hard cutoff by the diodes. P1 is a trimmer pot or an inline stereo volume control, such as those made by Koss or Radio Shack.

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Figure A10

The limiter can be set for only one headphone at a time. Different models of headphones have different sensitivity ratings, so the limiter must be readjusted if the headphones are changed. The more accurate and safest way to set the limiter is with an audio level meter and a headphone coupler (or artificial ear) sold by audiometric suppliers. If such equipment is not available, the limiter can be set by ear, but with less reliable results.

For initial testing, it is a good idea to use a pair of disposable headphones with the same impedance and the same or higher sensitivity as the intended headphones. Begin by turning the amp’s volume control to minimum. Do not connect the headphones yet. Feed an audio signal into the amp and turn up the volume until the diodes are forward-biased and clipping the signal. Use a voltmeter set on AC to confirm that there is about 0.7V across each of the diodes. The voltage should stay at about 0.7V even if the volume is turned up higher, indicating that the diodes are clamping the signal.

Set P1 in both channels for maximum resistance or set the inline volume control to minimum volume. With the trimmer pots, only one channel can be set at a time. With the inline control, both channels are set simultaneously, but if the channels don’t track precisely, always set the limiter based on the channel that is louder.

Connect the disposable headphones. Adjust P1 or the inline volume control slowly until the headphone volume reaches the desired level. Confirm that the limiter is working by turning up the amp’s volume control. If the volume increases, reduce the volume by readjusting P1 or the inline control. Repeat until the circuit clips at a consistent volume level. Then turn the amp’s volume control down to minimum and plug in the intended headphones. Slowly increase the volume and confirm that the clipping level is set correctly.

Once the pots are set, the settings must be protected against accidental change. While trimmer pots on a circuit board would be protected by the amp’s enclosure, it’s best to fix the thumbwheels in place with a dab of white glue. If an inline volume control is used, wrap the thumbwheel with electrical tape. For tips on setting maximum headphone volume, see Preventing Hearing Damage When Listening With Headphones. For more information on limiters, see Designing a Limiter for Headphone Amplifiers.

Addendum

12/4/98: Adding wiring diagram for headphone jack in figure 1.

11/25/98: Rewired SW1 in figure 1 to eliminate hum. Corrected R1 in figure 2.

11/20/98: Revised R1 in figure 1 to range from 100K ohms to 470K ohms, depending on desired input attenuation.

5/22/99:

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Jason Portman built the above version of the pocket headphone amplifier with an anodized aluminum case by Context Engineering, Inc. (available at Fry’s Electronics), volume control (10K) and blue LED. The larger size of the case allowed the use of 1uF WIMA polypropylene capacitors to couple the input. Very nice!

7/7/99: I have just been told that Digi-Key is backordered on the Burr-Brown opamps used in this project for the next 15-23 weeks! Here are some other sources: Insight Electronics and Sager Electronics. I have never order from these companies, but they are listed as Burr-Brown distributors.

7/12/99: Corrected polarity of LED in figure 2.

7/14/99: Added section on converting the pocket amp into a personal monitor.

8/24/99: Mika Vääräniemi built the modified Linkwitz acoustic simulator and pocket amp in a single aluminum enclosure. The power supply is an AC adapter that outputs 9VDC regulated. Here is the parts placement and wiring diagram that he used:

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He added a switch (S3) to turn off the treble boost and changed the values of C1 and R1 to C1 = 1uF and R1a = 50 ohms, R1b = 100 ohms. These values seem to give the widest soundstage with the least effect on the high frequencies. “[B]efore I was positioned in the middle of band playing music. Now I’m in the front row as close as you can be…. Music just sounds realistic and that’s what I was looking for.” A more complete description of his work can be found in the DIY Workshop Forum.

DIYers who would like to built both the simulator and amplifier together may want to scale the resistors and capacitors of the simulator section to increase the input impedance to about 2K ohms (x10 for resistor values, ÷10 for capacitor values – and use a volume pot between 10K and 50K ohms). Increasing the input impedance is not absolutely necessary, but it may then work better with some preamps which have a high output impedance.

8/25/99: Here are pictures of Mika Vääräniemi’s completed headphone amplifier with acoustic simulator:

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9/2/99: Jim Burruss built a “micro mixer” based on the pocket amp design. He used a metal candy box for a compact enclosure that also provides excellent shielding:

Attached are some digital photos of the micro mixer I built based on your design. I’m an electrical engineer and musician. I play a MIDI horn and needed a way to mix the signal from an electronic metronome with the output of the synthesizer for quiet practice. Your design ideas were great. I had an old Altoids box that looked just big enough to house it.

It has one mono input for the metronome with on-off and volume control on the pot with the short shaft. The other channel is stereo with its own ganged volume control. [Editor: The pots are available from Radio Shack.] The output is to drive headphones. I built it with an LM358 dual opamp just to verify the wiring and have an OPA (same pinout) on order to improve the sound.

The Altoids box provides great shielding. The board is insulated from the box with a fold-up plastic box made out of the packaging material from the metronome.

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9/7/99: This version of the pocket amp by Tomohiko Ishigami uses the acoustic simulator circuit by Jan Meier (see A DIY Headphone Amplifier With Natural Crossfeed). He reduced the gain of the amp to unity to minimize problems with noise, which he later traced to the CD player itself. The larger case is from Radio Shack (RS 270-213).

I feel it is very good idea to use modular approach. I used separate board for crossover and the buffer itself. This way, I did not have to go crazy load all the parts on one board which will result in a hay wire. Also, this approach is useful when I was trying to achieve smaller size.
I was able to use 1uF polymer capacitor for input…. These are so tiny. It is made by Phillips and you should be able to find it in Digikey [Digikey part nos. shown below]. I used this same type for my crossover circuit allowing me to conserve a lot of space:

3019PH-HD 1uF Metal Film Box ( 10mm (H) by 7mm (W) by 6mm (L) )
3015PH-ND .22uF Metal Film Box
3011PH-ND .047uF Metal Film Box

tomo4

11/21/99: Added section on replacing level switch with a volume control.

11/21/99: Stephen Jenkins wrote: Wow, I just finished building the headphone amplifer that you designed. I am in awe at the sound quality while using my little (but fabulous) Koss Porta Pro Jr’s and my Pansonic SL-S360 portable CD player. The only change I made was that I included an AC jack on the side so that I that I could plug into the wall while at home, this was really easy and I highly recommend it. Thank you for the plans, you’ve made my day!

12/18/99: Added section on implementing a dual 9V power supply for driving very high impedance headphones.

1/7/00: Several DIYers have installed Jan Meier’s natural crossfeed filter as a front-end to the pocket amp. Jan offers these tips re: selection and placement of a volume control for this combination: It all depends on the specific circuitry. Generally it might be better to place the pot after the filter instead in front of it. The influence of impedance changes might be less pronounced. A 10 kOhm pot will certainly be too small. 50 kOhm will be a kind of minimum I think. However, note that with certain opamps this will result in changing offset voltages, since the DC impedance changes with volume.

1/12/00: scrazy@gcn.net.tw built this version pocket amp, which has a 10K ohm volume control and an acoustic simulator front-end by Chester Simpson (see design by Fred Peng below). He used OPA134 opamps and set the gain to unity because his CD player’s line out supplies more than adequate drive voltage. Full details can be found at DIY Zone (in chinese only). His system consists of a Rega Planet CD Player and Audio Technica ATH-f15 headphones.

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1/13/00: Fred Peng’s headphone amplifier incorporates the acoustic simulator by Chester Simpson, except that he replaced the R4,R6 combination in Simpson’s circuit with a 100K ohm resistor and added a unity gain input buffer stage made from an OPA134 and a high current output stage made from a PMI BUF-03 buffer. The opamp power supply is double regulated for the cleanest output. The first stage of the power supply outputs ±34VDC, which is regulated to ±22VDC and again to ±15VDC. TWhen compared with a McCormack Micro Headphone Drive, the BUF-03 driving his Grado HP-1 headphones with the simulator bypassed sounded better in the high and low frequencies than the McCormack, but the McCormack was better in the mid frequencies. With the simulator switched in, the sound was more relaxed, the low frequencies were slightly “nasal”, and the soundfield moved from inside his head to outside. He is very satisfied with the result and is planning to make another simulator for his Stax Lambda headphones. Full details and schematics (in chinese only) can be found at DIY Zone.

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2/7/00: Eric Lee‘s pocket amp has a modified Linkwitz acoustic simulator front-end. He says “it works great…nice design…and I can hear almost no audible noise from it.” He used a slider pot for the volume control and installed dual headphone jacks for 1/4″ and 1/8″ headphone plugs. The enclosure is from Radio Shack.

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5/1/00: Forest Chang built a pocket amplifier with a modified Linkwitz simulator front-end and the component value changes suggested by Mika Vääräniemi (see above). He writes: The circuit that I built is the same as Mika’s, but the OPA that I used is the OPA134, and I put an OPA2134 as a buffer in the front-end of the acoustic simulator. The grounding method that I use is to tie the output ground, power supply virtual ground and switch housing together. Then I connect this common ground to touch the metal watch box (the enclosure that I used) with a spring. The amp has no hiss, even when I put it right beside the monitor. And I cut a beautiful picture from a metal candy box and put it into the watch box. My girlfriend uses the amp with a Panasonic SL-280 and Sennheiser HD-320 headphones. She is very happy with the sound improvement, and the cute headphone amp.

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5/1/00: Revised figure 2a and section on using dual 9V supply. Added section on constructing pocket amp with adjustable gain. Expanded description of how to cut protoboard with a utility knife. Added figure 6a – pocket amp with balanced input.

5/1/00: Jeff Medin‘s pocket amplifier has 3 sections: a gain stage, the crossfeed filter by Jan Meier and an output buffer stage. The power supply creates a virtual ground with a Texas Instruments TLE2426 voltage reference instead of a resistor divider network. The 1uF (or less) capacitors are Philips box-type metal film; capacitors larger than 1uF are Panasonic FC/Z series. All resistors are 1/4W Yaego metal film. Medin writes: This is the FIRST amp I built after discovering HeadWize. It is a “basic” pocket amp with the natural crossfeed circuit by Jan Meier. ALL parts are from Digikey. It has very good decoupling with 3 capacitors per opamp and 3.9uH chokes (the 4 green things that look like resistors – they are connected in series with each V+ and V- lead). The first stage (on the left side of the first picture) is an OPA2132 with a gain of 10.

This then feeds a Meier crossfeed circuit (4 caps in a row) and you can see the crossfeed resistor on TOP of the board (2.2k) with long leads. The output from the filter feeds a voltage follower (OPA2132) stage. The switches are for low and high crossfeed, power, and bypass for binaural recordings. I used Philips Box style metal poly caps. The two large caps on top & bottom of board are 1uF input caps. The output is taken from the OPA2132… with a 100 ohm resistor… which is included in the feedback loop so it will drive very low z phones and to prevent oscillation due to capacitance from long cables. I used 100 ohm resistors in BOTH stages.

If the resistor is OUTSIDE the loop, the impedance WILL have an effect on the sound of the phones, sometimes more bass, sometimes MUCH less signal based on the efficiency of the phones, etc. etc. Some phones as you know are spec’d to be run from an impedance of 100-150 ohms or so. I have a 15 year old APT/HOLMAN preamp (designed by same guy that invented THX-Tom Holman) and it’s Headphone Jack is driven by a 5532 with a 120 ohm resistor OUTSIDE loop right to the jack. I would suggest people can try both (like Jan did) and see what sounds better to them. I would DEFINITELY recommend that you include this resistor in at least the last stage.

Note that I did not have any problems, I always “over-build” opamp circuits so I don’t have to worry about problems later on. It’s just habit.

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5/4/00: Jasmin Levallois‘s amplifier is similar to Jeff Medin’s, except that he uses the Meier enhanced-bass natural crossfeed filter (and the original resistor-based virtually-grounded power supply). He writes: Finally I got some free time to complete my project…. I got a lot of work to do for school during the last few weeks and I didn’t have time to work on my amp. This weekend I decided to take one day to transfer the amp from the breadboard to the pc board. I used about the same circuit as Jeff Medin. The input stage has a gain of 10, the output stage is a voltage follower, and in the middle I put the Meier bass-enhanced crossfeed circuit.

I used 2 OPA2132 opamps, but if I had to do it again I would use 2 OPA2134. An OPA2132 costs $6.99 while an OPA2134 costs $2.67. Since there is almost no audible difference between both opamps, I would go with the OPA2134 to save money. Since the second stage has no voltage gain, I decided to omit the capacitor in front of the output stage. I also removed the resistor in front of the output stage, and I don’t hear any noise from the output stage. The only noise I can hear, sometimes, is coming from my CD player.

As you’ll see on the photos, the inside of my amp is very messy, but, hey, its my first electronic project. Fortunately, even if it’s messy inside, the outside looks pretty good. I really like this Serpac Enclosure (Digikey part no. SRH65-9VB-ND); it looks ways better than the PacTec case.

The photo of the battery compartment is to show that the Serpac enclosure has a 9v Battery compartment with battery contacts. It’s easier to remove the battery with that kind of battery compartment than the PacTec Enclosure. Also the Serpac enclosure is just about the same size as the Pactec enclosure except that it’s a bit longer, and the height is a little bit less. This might be a problem for the electrolytic capacitors. I would recommend the Philips ones with this enclosure rather than the Panasonic Z series because the Philips electrolytic caps are much smaller.

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Download parts list for Levallois Amplifier (MS Excel format)

6/16/00: Jasmin Levallois writes: This weekend I finished to build another pocket headphone amplifier for a project in my physic class. I used an Altoid box like Jim Burruss did, but I used the cinnamon kind to not be accused of plagiarism ;). This enclosure has the advantage that I could show to other students the circuit, and it is also very small and provides great shielding.

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I took my original circuit and I improved it a little bit. First, I replaced all my Phillips polymer capacitors by some Polyester made by Panasonic. I followed Jeff Medin’s recommendations and added a 100 ohms resistor in the feedback loop of the last stage. I used a .12uF capacitor to decouple each power supply pins. I also added a 100k resistor connected to the ground in front of the output stage. On my last circuit I had omitted this resistor, but many people in the forum convinced me to put it back.

I built the complete circuit on a very small board (4cm by 5cm) and I don’t think it would have been possible to make it much smaller than this. To save some space on the board, but also because Digikey was out of OPA2132, I used a single OPA4134. It is pretty cheap, $2.30, I think, and I really recommend it. I had a hard time to find some good electrolytic capacitors that would fit in the small enclosure. Finally I used some mini alum electrolytic capacitors made by Panasonic. You can find the Digikey part # of these capacitors in my part list.

This amp sounds great and looks great; I love it.

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Download parts list for Levallois Amplifier (MS Excel format)

10/11/00: Added sections on AC adapters, troubleshooting noise. Revised section on volume controls.

11/23/00: Bob Scott put his pocket amp into an Altoids candy box and uses it between his Sony MD player and his Sennheiser HD495 headphones. He writes: Attached are photos of my amp. I built it into an Altoids tin, partly for shielding, partly for the entertainment value. The only changes I made from your schematic was a slightly larger resistor for the LED to reduce current draw and using a “pigtail” for the input to save some panel space and reduce bulk when “cabled up”.

I got the short-handled switches from Digikey. They kept the unit compact and reduced the likelihood of the amplifier turning on accidently. I may build a second copy using “dead bug” construction to see if I can make it REALLY small.

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11/23/00: Carl Hansen has designed PC boards for the Levallois version of the pocket amp with Jan Meier’s enhanced-bass crossfeed. He writes: I have been spending the past nine months following your forums and building a number of variations of the pocket amp. I have decided that I have more than a few friends that would like to have one for Christmas in either kit or a variety of assembled form…. Because I have the resources available to me through my work I have gone ahead and laid out a nice little double sided board using Tango PCB, which I have sent to one of the commercial board houses in the Seattle area for a small “prototype” run. My boards arrived last week and I have assembled three of them and they work great!

The board house that fabricated the boards is fully automated meaning that no human hands were involved in the manufacturing process including a complete optical inspection using a robotic vision system…. I would like to sell off some of my excess boards. The price to sideliners in the forum like myself will be $6.50 each (or 3 for $17.00) plus $3.00 S&H; which is about the same as the cost for using Vectorboard. To those that have posted contributions to the forum that have furthered the dialogue, particularly regarding the pocket amp, I would like to offer two boards each for free except the cost for S&H.;

The specifications for the board are:

Dimensions: 1.80″ X 2.45″ with routed notches and corners to precision fit Pac-Tec case HML-9VB, leaving a 1.25″ space behind the panel for components such as switches and jacks. The amplifier section is designed for dual OPA2132/4s with the crossfeed filters between the amplifier sections. There are provisions for two levels of enhanced-bass crossfeed filters plus flat. A 3 pole, 3 position rotary switch or some equivalent would be required to use all three settings. The filter capacitor component locations have multi-holes each to allow the use of different size capacitors. There is a provision for volume control or high-pass filter resistor. Gain of course is a matter of component selection. Personally I have found a gain of 5 to be the most versatile. There is also a provision for power indicator LED.

Shown below are the Levallois schematic and pictures of the Hansen PC board. For more information about the circuit, see Levallois’ entry in the addendum update (p. 1) for May 4, 2000.

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Update: C.E. Hansen is no longer selling the PC boards or the Noble XVB93 mini-pots described in the article. Instead, Jon M. Tsukiji (JMT in the forums) is now selling the PC boards for the same price, although he is NOT selling the Noble pots. JMT is also selling completed amps in the Penguin Mints boxes first shown by “Apheared.” Contact JMT for pricing on the completed amps and to order the Hansen PC boards.

Jon M. Tsukiji
3142 Spruce Hill Ct.
Antelope, CA 95843
Email: JMT@surewest.net

3/14/2001: Major rewrite of article, including new appendix section on power supply options. Added new high resolution pictures.

3/14/2001: Coffin Lin put this amplifier (with a modified Linkwitz crossfeed front-end) in an old TV remote control case. He used an OPA627 opamp and made R2 and R3 in the Linkwitz filter adjustable instead of R1. The volume control is an Aiko pot in a shunted configuration with a 50K resistor (Dale RN55D), so that the audio signal passes through a single high quality resistor. Regarding his selection of the opamp, he writes:

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I found that the OPA637 oscillated, even though the gain was set to greater than 5. The power supply voltage was not symmetric (2V/10V using a 12VDC AC supply). Then I changed the opamp to an OPA627, which was quite good for both my Sennheiser and Pro2 headphones, but the supply voltage was still not symmetric enough (6.4V/6.8V). The OPA134 got best result in stability (6.5V/6.7V), but the sound is too fat for me. So the final version is OPA627 – great detail, sound balance, clear, dynamic.

clin2

Lin put the Linkwitz filter at the input to the amplifier. The component values in his version of the filter are:

R1: 30K ohms
R2a, R2b: 15K, 10K
R3a, R3b: 50K, 100K
R4: 33K
R5: 33K
C1: 3,300pF
C2: 10,000pF

The resistors are Dale RN55D. About making R2 and R3 adjustable, he says: I mistook R2 for R1, but on the Excel worksheet simulator, R2 can still alter some balance. I think that adjusting R3 is more effective than adjusting R2 (I forget which switch is for what resistor.) One has more stereo (good for dance and rock) and the other is more natural (good for jazz).

12/26/2001: Revised value of the current limiting resistor in figure A7. I reviewed Stephen Lafferty’s circuit for charging a single 9V NiMH battery. The value of current-limiting resistor in Lafferty’s circuit assumes that the specified unregulated 12VDC adapter will output 14VDC, because the amp is a very light load for the adapter. The recommended adapter in my project has a regulated output, so the output should be 12V exactly (or fairly close). Therefore, I changed the value of the resistor from 330 ohms to 220 ohms to get a charging current of about 20mA.

12/26/2001: Here are three candy box amps from forum members Doh, Droche and LivingPlasma. Doh put his Hansen-board amp in a Penguin Mints box (first shown by Michael Shelton – a.k.a. “Apheared”). He writes:

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It looks like Apheared beat me to posting a Penguin Mints amp, but I swear I didn’t steal the idea! Penguins rock! I’m afraid my amp isn’t nearly as DIY cool as Apheared’s creation, but it’s only my second amp and I just learned how to solder a few weeks ago!

As you can see, the Hansen board is mounted upside-down in the tin with the power and crossfeed switches sitting right underneath the op-amps. It has dual pigtails and one position of crossfeed plus flat. There’s no LED and volume is adjusted via an inline volume control from radio shack (soon to be replaced by a DIY version that uses the panasonic pot once those parts get in). I don’t have any of that fancy tape, so I actually just stick a metrocard under the lid before I close it. (Haven’t gotten around to glueing it in with some artist’s spray mount quite yet).

I think that there is still enough space in the box to wire the pot inside if anyone feels like giving it a try. Personally, I like the flexibility that a modular volume control gives me. On the other hand, I’m still trying to think up a way to get rid of the pigtails to improve portability.

Just a note on drilling the holes in an altoids tin or other metal candy container. What I found to work really well are the black and decker “bullet” tip drill bits. They have a small extension at the point that bites into whatever you’re drilling into so that the drill bit doesn’t slip. The tip works its way through the metal fairly quickly, so after it’s through you have a pilot hole that holds the bit steady while the rest of the bit does the work. Using these bits, I found drilling holes up to 1/4-inch to be no problem. The bits are available through amazon.com, but should be widely available.

Droche put his amp in the popular Altoids tin. He writes:

droche_amp.jpg

For any of you who think that building an amp is too difficult for a beginner- I am proof that it isn’t. I started browsing the forums a month ago with no electronics experience whatsoever. After browsing for a while, I put in a few orders and before I knew it, I had a headphone amp. It took a few tries to get it into the box, but I finally got it in after removing the headphone jacks and adding pigtails and removing the pot. It was well worth the effort. I was amazed at how much better the sound out of my portable MD player got. Thanks to everyone here for all the helpful info.

Livingplasma put his amp in a round candy tin. He writes:

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Not to take the attention away from Apheared, but I just couldn’t help it after seeing all thse proud people post their version of the CMoy pocket amp. Those who have been here a while will know I had a string of bad luck making my first cmoy, this is what I came up with the leftover parts. It’s the basic CMoy amp made with an OPA2134 and modified for the Meier crossfeed (changed some values so I could use a 50k pot and smaller input capacitor; yes, it’s unbuffered). Input is through the pigtail, has a LED power indicator and uses one submini toggle for power and one for the crossfeed (on or off). Measures about 3 inches in diameter (not counting the controls), and just under an inch in height. Schematically, I think it’s very similar to Tomo’s version.

Drilling the holes on the side of the tin is annoying, to say the least. After a certain hole size, it’s really hard to drill a hole, the bit catches on the metal and goes ripping the case apart (lesson learned trying it with a Altoids tin). I just made the hole as big as possible with the bit, then reamed it with either a screwdriver or a knife. The opening for the volume pot (it’s those panasonic ones) is a square, I think I used some old diagonal cutters I didn’t mind messing up and some pliers to bend and break the tin.

12/28/2001: Kenji Rikitake (a.k.a. “bdx” in the forums) built two versions of the pocket amp with an opamp-based virtual ground. He says:

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The amp on the left in the picture is the OPA2134 version; the one in the middle is a single-amp version (OPA134). The breadboard on the right is for the OPA2134 version. The basic amplifier circuitry is same, but I changed the value of the feedback resistors to 1k/4.7k ohms pair. The 1k ohm resistor at the + input of the opamp protects it from accidental overcurrent or overvoltage (though the probability is very low). This is generally recommended when you make a non-inverted amplifier.

bdx_amp3.gif

I tested with several different opamps. OPA2134 showed its excellence (only 3mV maximum output offset). NJM4580DD worked OK though it had 70mV maximum offset. The NJM082D (TL082 compatible) also worked but it could not fully drive my Sehnheiser HD414 Classic. Note that the NJM2043DD didn’t work (caused self oscillation).

bdx_amp1.gif

I also added an opamp voltage follower for providing the virtual ground, to stabilize the voltage exactly to the 1/2 of the unipolar power supply (namely a 9V dry battery). I tried the OPA134 as a unity gain buffer for the virtual ground driver and then the BUF634 (as ppl suggested). The quiscent current of the chip lowered from 4mA to 1.5mA or so, and the amplifier sounded the same. Note that OPA134 and BUF634 are virtually pin-compatible if you use the OPA134 for a unity-gain buffer. The 1N4002 diode protects the circuit from accidental inverse voltage connection.

The circuit is put into an aluminum case (T-SIN Denki TM-1) which can hold a 9V dry battery inside and can mount a 47mm x 72mm breadboard widely available here (Sanhayato ICB-88 compatible) in Japan. The FatBrain.com stickers are some of which I’ve got from the bookstore. The size of each amp is 87mm(W) * 31mm(H) * 103mm (D). I built the circuit on a glass-epoxy DIP breadboard. Since I proved the amp works fine with my Sony D-E880, Diamond Rio500, and Sehnheiser HD580 as well as with HD414Classic, I think I’ve got to build another one for my wife sooner or later. smile.gif

More details about Bdx’s amp and power supply can be found here.

2/21/2002: Added note about insulating the power jack ground when the power jack is mounted in a metal enclosure.

2/22/2002: Forum member Tangent has created a tutorial for electronics newbies who are interested in building the pocket amp. Many DIYers have found the tutorial very helpful. Please note that Tangent’s opinions are not necessarily the same as this author’s.

5/20/2008: Added a new appendix 1 about how to optimize the amp for use with low impedance headphones. Updated sections on choosing a volume control and using rechargeable batteries. Revised figures 1 and A7.

6/30/2008: Revised Appendix 4 and figures A8, A9 and A10.

Designing an Opamp Headphone Amplifier.

opamp

Because a few milliwatts will drive headphones to full volume, a great headphone amplifier design can be relatively simple. Yet, there are any number of reasons for experimenting with more complex topologies, such as improved performance and the ability to incorporate custom options. Of course, some DIYers like to try different circuits just for the fun of it. The major disadvantage of complex circuits is that they are complex. It can take a DIYer months to locate and purchase the parts, not to mention the time to assemble and troubleshoot the project.

Integrated circuit opamps are both complex and simple. They may contain hundreds of components on a chip, but are relatively easy to configure. For DIYers short on time and patience (and few people have the luxury of both), opamps are a convenient entre into the world of complex design. The audio cognescenti have attacked opamps as being one of the major causes of “mid-fi” sound, but if the truth be known, they are lurking everywhere – even under the covers of prestigious high-end gear. Opamps are not all the same. Building a headphone amplifier with good sound is a matter of careful selection and design.

This article discusses several opamp-based headphone amplifier circuits, including suggestions for selecting opamps, input coupling and filtering, high current output stages and power supply options. There are no recommendations for specific opamp brands or models. For tube devotees, there is also an introduction to designing with tube amp-blocks. Tube amp-blocks (AC feedback amplifiers and tube opamps) are not as compact as their silicon brethren, nor do they measure as well, but they do offer smooth tube sound with the ease of feedback configuration.

SELECTING SOLID STATE OPAMPS

Entire books cover the subject of interpreting opamp specifications. Here are a few guidelines for choosing opamps when designing headphone amplifiers. Opamps inch closer to the “ideal” with every succeeding generation. Modern devices are internally compensated for stability, have slew rates going through the roof and noise and distortion numbers at threshold of measurement. There are even opamps that will run off a 1-volt supply. For portable devices, the power supply requirements should be the first consideration. The majority of modern opamps will run with as little as ±4V, but low voltages may degrade performance. Check the manufacturer’s VCC specs to confirm that low voltage operation is, in fact, recommended. The most common battery supply voltages are ±1.5V, ±3V, ±4.5V and ±9V. Single supplies are another possibility. Keep in mind also that the idling current for the entire amplifier must also be low – around 10mA or less for good battery life. For more information, see the section on battery power options below.

Opamp performance specifications are an unreliable indicator of sound quality. So long as the numbers are below audibility thresholds, specs that are magnitudes better than the averages will not necessarily translate into better sound. Regardless of type (bipolar or FET), modern opamps do very well on the test bench. Total harmonic distortion figures are so low (typically less than 0.1%) that datasheets have stopped listing them. Look for noise specifications, listed as “noise density” in units of nV/Ö(Hz), of 25 or less, slew rates of 5uV/sec or more and “wide” unity gain-bandwidths of 3 MHz and higher.

opamp1a
Figure 1a

When reviewing the gain-bandwidth specification of a bipolar-input opamp, also examine the open-loop bandwidth. The gain-bandwidth defines the amount of small-signal gain at any frequency and is the product of the open-loop bandwidth and the open-loop gain. Most opamps have a high open-loop gain (100dB or more) and a relatively narrow open-loop bandwidth (100Hz or less). In a multi-stage system with overall feedback, if the opamp has a bipolar input stage and narrow open-loop bandwidth, it can manifest dynamic phase shifts and other response non-linearities with high level, high frequency input signals.

To reduce this type of distortion, choose a bipolar-input opamp with a wide open-loop bandwidth (into the kHz range) or use a FET-input opamp. FET input stages are more linear and so less susceptible to this type of distortion. Finally, the open-loop bandwidth of the voltage-gain input stage can be effectively extended with local feedback (see the section on output stages below).

Also look for unity gain stability and low offset voltage. Opamps that are internally compensated are less likely to oscillate at high frequencies, and save the builder the hassle of adding external compensation (however, it never hurts to check the amplifier output on an oscilloscope anyway.) The ideal opamp has zero DC output at idle, so that DC coupling can be accomplished without trimming. Real-world opamps have a small output voltage at idle. If the opamp is not followed by a gain stage, a 15mV or less offset at idle should be acceptable. FET-input opamps are known for their low offset voltages.

Are there audible differences between opamps with similar or identical specs? Some listeners can distinguish between products, but not all. Because modern opamps are internally compensated and are usually plug-in replacements for each other, building circuits with IC sockets or on protoboard first allows the DIYer to audition a variety of opamps at will. Theories abound as to why opamps may have sonic signatures in spite of stellar test results that suggest neutral sound. Years ago, IM, DIM, TIM, etc. distortion were held to be the culprits. Two of the most recent courses of research on this topic have been the effects of the harmonic structure of opamp noise and opamp input errors.

opamp1b
Figure 1b

The first course of research posits that much distortion in audio signals is actually noise, which may be too low to measure but is still audible. Human hearing is very sensitive to high order harmonics produced by high negative feedback ratios. Noise structures with a predominance of even order harmonics seem to sound less harsh. Opamp systems with poor harmonic structures can have improved performance if the systems are designed so that the harmonics cancel or harmonize with the products of other stages in the system. Figure 1b is a plot of the noise spectrum of a common bipolar-input opamp. Manufacturers generally do not include such analyses in datasheets. Since these tests must be done with sophisticated equipment that can measure noise 140dB or more below the signal, most DIYers will have to rely on other published sources for this type of data.

There are three types of opamp input errors that potentially affect sound quality: source-impedance, power-supply and thermal errors due to the output loading of an opamp. Source-impedance errors arise when there are unequal source impedances at each of the two inputs to an opamp, which interact with the opamp’s internal capacitances to create even-order harmonic distortion. It is a common-mode type error, and so applies only when the opamp operates in a non-inverting configuration. JFET-input opamps have an internal capacitor at each of the inputs, and are likely to show higher levels of source-impedance distortion than bipolar-input types.

Source-impedance errors can measured by comparing distortion levels when the feedback network impedance (Rf||R) differs from the input source impedance Rs and when they are the same. Selecting opamps with low internal capacitance or balancing the source impedances will minimize this form of distortion. The latter technique is discussed in the section on configuring opamps for voltage gain below.

opamp1c
Figure 1c

Power-supply errors occur when noise from the power supply mixes with the input signal. The PSR (power supply rejection) specification is a measure of how well an opamp is able to block power supply noise and values of 100dB or more are common. PSR will vary with frequency, but the spec usually refers only to DC behavior. Instead, search for a graph in the datasheets of the PSR over the audio frequency range. In addition to choosing opamps with high PSR over a wide audio range, power-supply errors can be reduced by using power supplies that are highly regulated and bypassed.

opamp1d
Figure 1d

Opamp-based headphone amplifiers can be prone to thermal errors due to output loading of the opamps. The power dissipation in an opamp when driving low impedance loads can raise the temperature of the device and cause changes in the input offset voltage, thus compromising the linearity. In dual and quad IC devices, the thermal conditions of one opamp can affect all others in the package because the opamp circuits share a common substrate (“power-dissipation-related crosstalk”).

Thermal errors can be measured by comparing the output distortion of an opamp under load and no-load conditions. Figure 1d shows the thermal loading effect of one buffer on the other in a dual buffer IC (the system circuit is similar to figure 5d). Channel A is being fed a frequency sweep signal; channel B is idle. When buffer A is driving a 25 ohm load (as opposed to driving no load), it induces a stronger thermal-related error signal at the input of buffer B.

When the dual (or quad) IC buffers are used in circuits with voltage gain front-end as seen in figure 5d, these thermal errors can be corrected via global feedback. Conversely, buffering an input-stage opamp can reduce thermal errors in the input stage by isolating the power dissipation to the output stages. (For more information about buffering, see the section on output stages below.) In general, using single opamps instead of duals or quads will prevent power-dissipation crosstalk distortion. Precision low-noise opamps appear to have the lowest thermal errors.

TUBE-BASED OPAMPS

Unlike transistor amplifier design, tube amplifier design is more dependent on the electrical characteristics of the tubes themselves. Tube opamps attempt to bring the simplicity and higher preformance of amp-block design to tube audio. In audio applications, they can aspire to the same high performance as their solid state cousins and have the additional benefit of even-order distortion harmonics. There has been a revival of interest in these devices with the publication in recent years of several amp-block circuits, ranging from basic AC feedback amplifiers to tube-MOSFET hybrids – all configurable with the familiar opamp feedback scheme.

Not widely available back in the heyday of glass audio, tube opamps are very hard to find today. The following circuits develop the tube amp-block concept with increasing complexity. They all have limited current output and may need an output buffer stage to comfortably drive headphones (see the section on output stages below). To adjust the closed-loop gain of any of these amp-blocks, simply add a feeback resistor from the output to the inverting input and an input resistor – just as with solid state opamps. Some of these amp-blocks may prefer higher magnitudes of resistance than are typical of solid state opamps, so sample gain resistances are included. It’s a good idea to construct several of these at a time to have a handy supply for experimentation.

opamp2
Figure 2

Eric Barbour’s “1+1 Cascade” amp-block is an AC feedback amplifier (figure 2), consisting of a common cathode gain stage and a cathode follower output stage. This amp-block has a single inverting input and limited open loop gain, but is entirely suitable as a front-end of a headphone amplifier. The open loop figures are less than spectacular: G ~ -50, Fh ~ 30 kHz, THD > 2%, Rout ~ 2K ohms. When configured for a closed loop gain of -10 (Rf = 100K ohms, Rin = 10K ohms), the situation changes dramatically: Fh > 100 kHz, Rout ~ 500 ohms and THD drops below 0.4%. Since the 12AX7 is a dual triode, this design uses only 1 tube per channel. 12AU7s can also be subsituted, but the open loop gain will be lower. Purists may want to add an inverting output stage for correct output phase.

opamp3
Figure 3

Fred Forssell’s circuit (figure 3) has the differential input stage of a true opamp with a high common-mode rejection ratio (CMRR) and a mu follower output stage (biased at 12mA). The open loop gain is about 510 (30 from the first stage, 17 from the second). When configured for a closed loop gain of 18, the performance approaches that of solid state opamps: THD < 0.1%, Rout= 8 ohms, Fh > 400 kHz and s/n ratio = -86dB. Despite the low output impedance, the load impedance should be 3K ohms or greater to avoid increased distortion. When configuring this opamp for gains of less than 18, Forssell recommends using a lower mu input tube such as the 12AU7A for a lower open loop gain, so that less feedback is required.

opamp4
Figure 4

Both of the Barbour and Forssell amp-blocks use high voltage supplies, and neither is DC-coupled. Erno Borbely’s hybrid design (figure 4) is both low voltage and DC-coupled. The differential input stage uses a single ECC86/6GM8 dual triode, which has a maximum anode voltage of 25V (a good substitute is the 6DJ8/ECC88). The current mirror Q1 and the constant current diodes (D1A and D1B) increase the CMRR and improve linearity. The output stage is a P-channel MOSFET configured as a common source amplifier with Q3 as its current source (the bias current is 10mA and can be adjusted by varying Rs). Rp is adjusted for 0 output voltage.

C2 provides phase compensation and if the opamp is configured for less than 6dB of gain, the R15-C5 low pass network must be added for stability (for G = 6dB, C5 = 100pF; for G = unity, C5 = 330pF). The open loop characteristics of the Borbely hybrid are excellent, especially for a tube opamp: G ~ 53dB, Fh ~ 90kHz and THD < 1%. When set to a gain of 10 (Rf = 10K ohms, Rin= 1.1K ohms), the specs once again are excellent: Fh > 700 kHz, THD < 0.1% and the output impedance is 50 ohms. A high load impedance (10K ohms) is recommended for maximum voltage output (15V).

CONFIGURING OPAMPS FOR VOLTAGE GAIN

opamp5
Figure 5

Opamps are most commonly used as voltage gain stages. The basic voltage-gain configurations are shown in figures 5a, 5b. The input impedance is the value of the input resistor. The output impedance Zo depends on the particular opamp, but generally decreases with decreasing gain (see the opamp datasheet for output impedance specs). If the opamp will be driving headphones directly, the output impedance should be less than 1/10th the headphone impedance across the audio spectrum. When choosing between inverting or non-inverting stage, the goal to keep in mind is that the opamp’s contribution should result in correct phase at the amplifier output. As a rule of thumb, non-inverting configurations tend to have lower noise, higher input impedance and wider bandwidth, but may be subject to certain design constraints (see manufacturer specifications).

Headphone amplifiers are usually fed from the outputs of a preamp or portable stereos which have plenty of voltage gain (instead, they lack the current capability to drive headphones cleanly). If a headphone amplifier has a voltage gain stage, the gain is typically set between 2 and 10. Some opamps sound cleaner at lower gains. The feedback resistor Rf probably should be less than 1M for optimal stability (check manufacturer specs for other feedback network design issues), and lower feedback network impedances (Rf||Rin) result in lower noise.

opamp5c
Figure 5c

Modern opamps do just fine with the basic configurations, but there are many design tweaks that can improve performance. One such optimization reduces source-impedance input errors in JFET-input opamps, which were discussed in the section on selecting opamps above. Recall that source-impedance input errors affect non-inverting gain configurations only and are caused by unequal source impedances at the + and – opamps inputs. The non-inverting amplifier in figure 5c balances the source impedances by choosing Rs = Rf||R. In a headphone amplifier, Rs is likely to be variable, in the form of a volume control, and so the 2K to 3K value is an approximation.

opamp5d.gif
Figure 5d

In a multi-stage opamp system (such as a voltage gain stage followed by a current buffer – see the section on output stages for more information), if the input stage opamp has a bipolar input stage and narrow open-loop bandwidth, it may exhibit nonlinearities when fed high level, high frequency signals. The system in figure 5d has an input stage opamp, which has had its open-loop bandwidth effectively extended under local feedback. The overall gain of the system is 5, but the local gain of the input stage is about 100 for an effective open-loop bandwidth of 100kHz. The bandwidth extension should go well beyond the audio range.

opamp6.gif
Figure 6

If the opamp is configured for a gain of 1 (R = Rf), it becomes a voltage follower. Most solid state opamps will also function as non-inverting followers with a straight wire in place of the feedback resistor (figure 6). Non-inverting voltage followers have the input impedance and a low output impedance. The input impedance of an inverting follower is the resistance of the input resistor. Voltage followers are often used as buffers which could drive headphones, but voltage gain opamps have modest current capability. A high current buffer opamp is specially designed to provide large amounts of current – perfect for driving headphones. For more information, see the section on output stages below

Handling Balanced Inputs

opamp7
Figure 7

Pro audio equipment may have balanced inputs and outputs – where the ground is separate from the signal ground for more effective noise shielding. Thus, each channel has a total of 3 connections: signal, signal ground and ground. The circuit in figure 7 converts a balanced input into a single-ended signal with unity gain (the input resistors are split to implement a RF filter – see below). The resistors must be matched to within 0.1% or the CMRR will degrade (e.g., an 80dB CMRR can drop to 60dB due to input resistor mismatch). The converter can also be configured with gain determined by the ratio of Rf / R, but keeping all Rs that same value makes matching the resistor array easier.

AC Coupling and RF Input Filters

Bandwidth-limiting the signal input can block DC voltages or filter out RF noise. DC protection is not necessary if the audio source already has 0 DC output, but some designers prefer extra insurance. With values of 1uF and 100K, the high-pass input filter in figure 5a has a corner frequency of about 1.6Hz and will minimally affect bass response or overall sound quality – if a high quality parts are used (e.g., film capacitors and metal film resistors). Instead of the resistor, an audio taper potentiometer could be substituted to serve as a volume control.

If the signal has RF noise, it can be cleaned up with a low-pass filter at the inputs. The low-pass network in figure 7a has a corner frequency of about 200kHz. An alternative RF network is shown in figure 7b. Frequencies above the corner frequency are mixed together, so that they are canceled out by the opamp’s CMRR. As with the resistor array, the RF capacitors should also be matched as closely as possible. Also use shielded cable when wiring the inputs to further reduce noise pickup.

opamp8
Figure 8

Low and high pass filters can be cascaded at the input, so long as the resistor values of each filter are different by at least a factor of 10. Also, the impedance of input network will affect the overall impedance of the input stage, so must be accounted for in selecting filter resistance values. These bandpass filters can alternatively be incorporated into the feedback loop. The circuit in figure 8 has an approximate bandpass from 2Hz to 150kHz. Some audiophiles may be able to hear distortion from capacitors in the signal path. As with opamp distortion, capacitor distortion is not audible to everyone. Before discarding the benefits, audition the amp on a protoboard with and without the capacitors.

THE OUTPUT STAGE

opa132_amp

Voltage-gain opamps may output enough current to drive some headphones directly (check the manufacturer specs). For example, the author built a pocket headphone amp (shown above) with Burr-Brown OPA132 opamps in a non-inverting configuration as shown in figure 5. The amp has no trouble reaching ear-splitting volumes with most headphones. For more information about this project, see A Pocket Headphone Amplifier. Modern dynamic headphones will play loudly with just a few milliwatts (see Understanding Headphone Power Requirements).

However, when an opamp does not have enough current capability or if it is susceptible to output loading errors (for information on output loading errors, see the section on selecting opamps above), it must then be augmented by an output stage. This section reviews solid state and tube class A followers, class AB symmetric emitter followers and buffer opamps (which are nothing more than elaborate emitter followers) as output stages. For tube amp-block front-ends, there is a discussion on interfacing tubes to solid state output stages. Note: the class A followers and high current buffers described below also make excellent standalone headphone amplifiers, where voltage gain is not necessary.

Class A MOSFET Follower

opamp9a
Figure 9

Purists prize class A amplifiers as capable of reproducing audio signals with the ultimate fidelity, because the output voltage swing is under the control of a single transistor or tube. Class A amplifiers are inefficient, consuming up to 400% more power than they output, but are enjoying revived popularity for their simple topologies (especially single-ended class A amps). Whereas class A loudspeaker amps run hot enough to heat a room, headphone fans can indulge without guilt, since headphones require very little power.

The MOSFET source follower in figure 9a is a single-ended class A output stage. MOSFET followers (and their bipolar cousins, emitter followers) are current amplifiers, which have non-inverting unity (or slightly less than unity) gain. A voltage divider biases the MOSFET. The bias pot Rp adjusts the output voltage to 0V for DC coupling (see Greg Szekeres’ Class A Headphone Driver for an AC coupled design). An input coupling capacitor blocks incoming DC and isolates the MOSFET’s bias network. Rs sets the MOSFET’s bias current ID. Then Rs = V/ID.

The MOSFET can be any power MOSFET, so long as the voltage and current ratings are adequate. MOSFETs have the “soft” overload characteristics of vacuum tubes and are preferred in this type of application over bipolars. The gate resistor helps to stabilize the MOSFET. The VDS spec should be at least twice the idle voltage. If the MOSFET idles at 1/2 the total supply voltage, then VDS should be at least the value of the total supply or higher. Rs is a power resistor. Starting with an idle current Id of about 100mA and -V = -12V, then RS = 12/.1 = 120 ohms. The resistor’s power rating should be much greater than 12 * 0.1 = 1.8W (at least 3.6W to be safe). Also make sure that the MOSFET is heatsinked to dissipate a similar amount of power.

The amp in figure 9b (by PRR) adds an opamp gain stage. The design exploits the ability of an opamp to serve as a voltage source. Once Id is set via Rs, the MOSFET gate draws the bias voltage it needs directly from the opamp’s output without a biasing network. (The MOSFET’s device specs will have a graph of gate voltage VG vs. idling current ID.) The feedback network (5K and 1K resistors) sets the overall gain to 5 and automatically nulls the output to 0V for DC coupling. That 20pF capacitor rolls off the frequency response above 100kHz to prevent oscillations. With Rs = 100 ohms, the MOSFET idling current ID is 120mA.

mos_cs.gif

Instead of a resistor for RS, a precision current source would improve linearity. Current sources are usually made with a transistor, but the version above employs a LM117/317 floating regulator, which needs only one resistor to adjust current output from 10mA to 1.5A. The voltage differential between Vin and Vout (which is the 1.25V internal reference voltage) should be between 7 and 15V. At higher differentials, the current output starts dropping due to internal safe-area protection, in which case more than one current source can be paralleled for higher output. While not required, the output capacitor helps eliminate any instability. Again, heatsinking is recommended.

AC-Coupled Cathode Follower

opamp10
Figure 10

From a design point of view, tubes are less favored as output stages, because the output impedance is higher than can usually be achieved with transistors. Yet, there are many excellent headphone amps with tube outputs. The AC-coupled cathode follower in figure 10 (from Andrea Ciuffoli’s headphone amp project) achieves a relatively low output resistance of about 33 ohms by paralleling two sections of a dual triode. The cathode resistor is tapped to provide self-bias. Each section is biased at 26mA or 52mA total.

The output impedance of a single-tube cathode follower is calculated as: Zout = Rk / (1 + GmRk), where Gm is the tube’s transconductance and Rk is the total resistance of the cathode resistors. Therefore, when building a cathode follower, select tubes with a high transconductance to get the lowest output impedance.

Class B and AB Symmetric Emitter Followers

opamp12
Figure 11

The high power consumption of class A amplifiers makes them impractical in battery-powered headphone amplifiers. The current booster circuits in figure 11 have complementary output devices that each reproduce one half of the audio signal. These schemes are more efficient because the idle current can be very low or even 0mA. The circuit in figure 11a is a class B amplifier with Q1 and Q2 off at turned off at idle. When the audio signal is positive, Q1 conducts; when it is negative, Q2 conducts. However, both transistors conduct only when the signal exceeds the forward bias voltages which is around 0.7V. Therefore, both transistors remain off when the audio signal is between ±0.7V, resulting in crossover distortion at the output. Since headphones are driven at low output voltages, this type of distortion is particularly noticeable in a headphone amplifier.

The circuit in figure 11b improves performance by allow the opamp stage to supply current until the voltage drop across R is large enough to forward bias both transistors. However, this design suffers from fluctuating output impedance. The output stage in figure 11c solves both problems by having both transistors conducting at very low idle currents. The voltage drop across the two diodes forward biases Q1 and Q2; the emitter resistors determine the idle current – about 0.6mA with these values. The output stage operates in class A at low levels – until the load draws more current or voltage swing than one of the transistors can provide. For battery operation, the output stage is often biased from 1-10mA, trading off between sound quality and battery life. The minimum idle current is best determined by monitoring a sine wave output on an oscilloscpe while adjusting the bias until the crossover distortion just disappears. AC powered amplifiers can take advantage of extended class A operation by increasing the bias current. Earle Eaton’s headphone amplifier uses a variation of this design. Sheldon Stokes’ headphone amplifier has a class AB MOSFET output stage.

High Current Buffers

opamp12
Figure 12

High current buffers are basically output stages on a chip. Because they are specialty products and are meant for use in specific applications, buffers are optimized to be particularly good at one job. In general, these chips have fantastic specs: slew rates in the hundreds, low distortion and of course, high current capability. For a headphone amplifier, a buffer that can output 100mA is probably more than sufficient, but additional current drive doesn’t hurt, so long as the power supply requirements meet the builder’s goals. Figure 12a shows a voltage gain opamp with its current capability doubled with the help of an identical opamp configured as a voltage follower. The load balancing resistors ( Rc ) are about 50 ohms. The output impedance of this circuit would be Rc || Rc || ( R + Rf ), but the impedance seen by the headphones is much less – reduced by the effect of the feedback taken at the outputs of the combined Rcs: Zout = Rout / amount of feedback.

Figures 12b and 12c show voltage gain opamps augmented with current buffers – 12b has a buffer outside the feedback loop and 12c has buffers inside. The overall gain for both versions is the same, but the version with global feedback might function with greater linearity. However, some designers argue that these buffers are already very linear, and global feedback can introduce instability into a system. Both configurations work. If the circuit of figure 12c is wired for local feedback only, such as in figure 12b, then the load balancing resistors can be as little as 1 ohm for a lower output impedance. When using dual or quad buffer ICs, global feedback can help correct for output loading errors (see the sections on selecting opamps and configuring opamp voltage gain stages for more information about output loading errors).

Note: Class A output stages can similarly be excluded from the feedback loop, but class AB stages should be included, since they are more prone to nonlinear operation.

In the case where a single buffer does not supply enough current or has an output impedance that is too high, it is possible to parallel output buffers. Figure 12c doubles output current capability and cuts output impedance in half by paralleling 2 output buffers. The current-summing output resistors Rc (typically 50 ohms) ensure that all of the buffers contribute equally to the output. Again, because the feedback is taken after the Rcs, the output impedance seen by the headphones is less than 1 ohm. Ben Duncan’s PHONES-01 headphone amplifier substitutes ferrite beads and incandescent lamps (see below) for the output resistors, reasoning that any unequal sharing is likely to be in the RF range. The beads also help block RF. Again, the feedback loop can be placed either before or after the parallel buffers.

Interfacing Tubes To Solid State Output Stages

When interfacing tubes with solid state output stages, the higher operating voltages of tubes pose two potential problems. First, the power supply may have to be “stepped down” and second, tube circuits can send out high voltage transients that could damage solid state components. The solutions: use high voltage opamps and buffers and/or limit the voltage going into solid state inputs. With a high-voltage MOSFET, the class A source follower described above would interface well with tube gain stages, as tubes and MOSFETs have similar sonic characteristics. The MOSFET amp has zener protection against overvoltage damage. There are also high-voltage bipolar devices, but they are less common. Apex Microtechnologies and Burr Brown are two manufacturers of high voltage opamps and buffers. Many of these are well-suited for audio applications, and a few chips are able thrive on power supplies of up to ±600V.

opamp13
Figure 13

High voltage output stages may also have input voltage limitations that tubes could breach. The following are two overvoltage protection schemes that can be used with any solid state output stage. Figure 13a is a suggestion by Eric Barbour. When fed high voltage transients, the zeners clamp the input to a maximum of ±15V. Figure 13b is the protection scheme that Greg Szekeres uses in his MOSFET headphone driver. Here, transients in excess of the power supply voltages will forward bias the silicon diodes and be conducted out of the system. The input resistor sets the minimum load impedance seen by the tube output.

Output Current Limiting

opamp14
Figure 14

When a headphone plug is inserted or removed from the jack, the possibility arises that the amplifier outputs will be shorted, if only briefly. Without current limiting, such a short could burn out opamps and/or output stage transistors. Rather than resort to complex current sensing schemes, figure 14 shows two common limiting mechanisms that protect against short-circuit damage: current limiting resistors and incandescent bulbs. Current limiting resistors set the minimum load that the amplifier can see – typically 100 ohms, 1/2W. Output resistors will reduce the output power and increase the amplifier’s output resistance, but most headphones will be unaffected. Another option is to locate the current limiting resistors inside the feedback loop (figure 12) so that the effective output impedance of the amplifier is minimized from the feedback. See Headphone FAQs for more information about the impact of amplifier output impedance on headphone sound.

In place of a current limiting resistor, an incandescent lamp has the advantage of very low resistance when the filament is cold. Lamp filaments have a positive temperature coefficient. As increasing current heats the filament, the resistance also goes up, thereby reducing the output current. Choose lamps with voltage and current characteristics similar to that of the output stage. Incandescent lamps were once popularly deployed to protect loudspeakers from overdrive. The idea resurfaced as output limiting for headphone amplifiers in Ben Duncan’s PHONES-01 headphone amplifier project.

EQUALIZATION

opamp15
Figure 15

Designing an equalization stage is an entire subject by itself (see Designing a Pocket Equalizer for Headphones). Equalization can be implemented in separate circuit blocks – either as active stages or passive networks – to ensure that they can be switched out completely without compromising the quality of the main gain stage. But there are instances where equalization is so important and basic to the use of the amplifier that the EQ filter network is incorporated in the feedback loop of the main gain stage for convenience and economy. For example, headphone amplifiers for guitar practice almost always require a bass boost.

Figure 15 shows a bass boost feedback network by T. Giesberts that gives a 10dB boost at 50 Hz when turned on. The network is a shelving EQ. With the boost deactivated, R1-C1 and R2-R3-C2 form a bandpass with threshold frequencies of about 20Hz and 30kHz. The gain of the amplifier is determined by (R2 || R3)/R1 and is approximately 4 with the values shown. With the boost switched in, R3-C3 create a bass shelf, with a threshold frequency of about 500Hz. The downturn in the low frequency response below 50Hz is caused by the attenuation from the input high pass filter.

ACOUSTIC SIMULATION

opamp16
Figure 16

Headphone sound suffers from a “super-stereo” effect caused by the isolation of each audio channel to one ear. Acoustic simulators electronically alter the stereo signal to create a more natural soundfield in headphones. They may be implemented with digital or analog filters (also called crossfeed filters). While digital and active analog simulators have amplification for headphones built into the design, passive simulators are RC networks that shape and time delay the crossfeed. Passive networks are sensitive to the source and load impedances that can affect the frequency response of the networks. (For examples of passive acoustic simulators, see the HeadWize Projects Library. For more information about digital and active network simulators, see Technologies for Surround Sound Presentation in Headphones.)

Depending on the input and output impedances of a passive simulator, it can appear at the input or output of a headphone amplifier (figures 16a, 16b), but isolating the network between two amplifier stages will often result in the best performance (figure 16c). With two isolating stages, the network can be assured of seeing a low input source impedance and a high output load impedance, such that the frequency response of the network remains constant. Both stages can be voltage gain blocks and/or unity-gain buffers, as the application may require.

However, with battery-powered amplifiers, which may operate the opamps at lower voltages, the preferred way to construct a headphone amplifier with an acoustic simulation is to make the second stage a voltage gain block to compensate for any insertion loss through the network as well as provide for overall voltage gain. If the voltage-gain block does not output sufficient current to drive headphones, add a high current, unity-gain buffer after the voltage-gain block.

When using multiple opamp gain stages, be sure to check the idle voltage at the output of the last stage. If it is more than a few millivolts, the DC-offset voltages of the opamps must be adjusted – either by trimming the DC offsets, by adding capacitors between stages and at the output to block offsets or by selecting feedback resistors to minimize offsets (see next section).

ADJUSTING DC-OFFSET VOLTAGES IN MULTI-STAGE AMPLIFIERS

opamp17
Figure 17

In single stage amplifiers, the opamp’s DC offset voltage is only a few milliamps and is rarely a problem. In multistage amplifiers, DC offsets may be amplified by successive stages until the idle voltage at the output of the final stage reaches several volts, although the overall gain the system may not be very high. Jan Meier experienced this situation while building and testing a headphone amplifier:

Referring to figure 17a, the non-inverted input of an opamp wired as a voltage follower requires a small input bias current (i+) that, since it flows through the resistor R1, generates a non-zero voltage V+ = (i+)*R1 at the input. Typical values for i+ are 1uA to 2uA (LM6171/LM6181/LTC1206) for bipolar-input, or 1 to 50 pA (OPA627/OPA604) for FET-input amplifiers. With a R1 of 100K, V+ (and thus Vout) can have values up to 200 millivolts!

In figure 17b, a feedback loop is added that amplifies V+ by a factor (R3+R2)/R2. It is not unusual for a headphone amplifier to have a gain factor of around 5. This will, however, also amplify V+ for a Vout of up to 1000mV, which can damage headphones – especially low impedance headphones. Fortunately the inverted input also generates a bias current (i-) that generates a DC-voltage (V-) at the inverted terminal and thus counteracts the effects of V+. The effective resistance to ground seen by the inverted input is the value of R2 and R3 in parallel which equals (R2R3)/(R2+R3). To eliminate the output voltage offset generated by i+, the input voltage V- should be equal to V+:

(i+)R1 = (i-)*(R2R3)/(R2+R3)To select values for R2 and R3, first take a look at the specifications of the opamp for i+ and i-. Note that they do not have to have the same value. For instance, the LTC1206 has an i+ value of 2uA whereas i- goes up to 10uA! By a proper selection of the resistor values, the offset can be strongly reduced. With a headphone amplifier made from a LM6171 opamp and having R1 = 47 kOhm, R2 = 56 kOhm, R3 = 300 kOhm, one channel shows a very good offset of only 20 mV. The other channel came down to a hardly measurable 0.2 mV! The fact that the channels were not equal simply has to do with manufacturing variations in opamps of the same type.

A problem remains with the input stage. If the input potentiometer is directly coupled to the opamp, the value of R1 now changes with the volume control, and a perfect fit of the resistances can not be made. A possible solution is shown in figure 17c. The resistance of the potentiometer no longer has an influence on the DC-resistance of the opamp. Alternatively, if the headphone amplifier has a second stage, the input stage can be decoupled from the second stage as shown in figure 17d. If the output of a first stage is directly coupled to the input of a second stage, the effective value of R1 is zero and a match can not be made. However, you simply can put a resistor between output and input.

Last warning: If the headphone amplifier will also be a preamplifier, any DC-offset at the output will be amplified by the power amp and will be fed into a low resistance loudspeaker. In this situation a few millivolts offset can damage the loudspeaker. To prevent any damage to loudspeakers or to the power amp, always use (decent quality) capacitors at the output of the preamp.

HEADPHONE DISTRIBUTION AMPLIFIERS

opamp17a
Figure 18a

Multitrack recording allows musicians to record songs in layers. Tracks can be added or overdubbed. Musicians may be positioned far apart from each other or play at different times to isolate their performances for the greatest flexibility in editing. Headphone monitoring is the most common way for musicians to hear each other under these circumstances, and a headphone distribution amplifier is central to this function.

Headphone distribution amplifiers can drive several pairs of headphones from a single set of inputs. While it is fairly easy to build one from a power amplifier with a ladder of output resistors (see the headphone FAQs for instructions), there are advantages to driving each headphone from its own amplifier, such as greater control over gain. The first stage of the basic distribution amplifier shown in figure 18a is a voltage follower that provides impedance buffering and signal inversion for correct phase at the headphone output. The buffer feeds any number of headphone amplifier blocks with their own volume controls.

opamp17b.gif
Figure 18b

As more musicians demand custom mixes, so commercial distribution amplifiers have begun adding mixer features. Figure 18b shows how to convert the input buffer stage of the basic distribution amplifier into a mixer stage. The input buffers (A1 and A2 for the left and right channels) now have a series of 100K summing resistors, one resistor for each stereo or mono input. The level controls for the stereo and mono inputs are balance-volume and pan-volume sets of pots. To move the balance or panning characteristic closer to the ends of the pot rotation, decrease the value of Ri.

opamp17a
Figure 18c

A full-featured headphone distribution amplifier will have limiters and possibly equalization stages for each headphone output. An acoustic simulation network, equalizer and/or limiter can be placed between the buffer and headphone amplifier blocks (see Designing a Limiter for Headphone Amplifiers for information about limiters and Designing a Pocket Equalizer for Headphones for equalization schematics). To increase the drive capability of an amp block, add a current buffer output stage (after any active EQ stages).

POWER SUPPLIES

AC Power Supplies

opamp18
Figure 19

To regulate or not to regulate, the answer depends on the circuit. Modern opamps have excellent power supply rejection ratios (PSRR) and are less affected by voltage fluctuations than older products, but discrete output stages may be more vulnerable. Since headphone amplifiers draw so little power and 3-pin regulators are cheap, it cannot hurt to have a regulated supply. Figure 19 shows a dual supply with the LM150/LM133 floating regulators configured for slow-start to minimize turn-on thumps. The delay is R*C = 8900 * 1000 E-6 = 9 seconds. Also check for power supply schematics in HeadWize Projects articles or in the datasheets for regulators. In any case, each opamp should be decoupled from the power supply with a 0.1uF ceramic capacitor and possibly a 10uF electrolytic, connected from the power supply pins to ground (see figure 14e below).

opamp19
Figure 20

Tube circuits often do not use regulated supplies, but where recommended, it is usually a single high voltage regulated supply for a gain stage and/or a low voltage regulated supply for tube heaters. For example, the Forssell tube opamp requires a regulated +350V supply for the output stage (at least 30mA for two opamps) and a 6V regulated supply for the heaters to minimize hum. Floating regulators, such as the LM150, can output hundreds of volts, so long as the input/output differential voltage remains within spec (and don’t forget to diode protect the regulator). Figure 20 shows a simple zener-based high voltage regulator. If the output voltage goes up, the potential across VGS decreases and the MOSFET reduces output current. If the output voltage goes down, then VGS increases, and output current increases. The IRF420 specified has a VDS of 450V and an ID of 2A continuous and must be mounted on a heatsink. The zener can be any series of 5W zeners (for the Forssell circuit) that total about 350V.

Battery Supplies

eq12.gif
Figure 21

There are opamps that will operate on a single 1.5V cell. Such micro-power opamps can drive very efficient, low-impedance headphones. With other headphones, the inability of micro-power opamps to develop higher voltages across the load will limit the volume. One method of getting higher voltages from batteries is to stack the batteries in series. Another is to raise the battery voltage with a DC-to-DC converter (to several volts or even several hundred volts in the case of portable electrostatic headphones). DC-to-DC converters, also called switching regulators, do their magic by changing the DC voltage to an AC voltage via an oscillator, which feeds a step-up transformer or capacitive/inductive reservoir to build the voltage, and then is converted back to DC at the higher voltage. As with any AC-based supply, a DC-to-DC supply must have a good filter network at the output to minimize power supply noise.

Opamps can run off single supplies, but are designed for dual supplies. Headphone amps with direct-coupled outputs must be powered from dual supplies. If there is room in the amplifier enclosure, separate batteries for the positive and negative supplies is the suggested implementation (figures 21b). If the opamp can run on a supply of ±3V or less, a single 9V battery can be converted into a dual supply as shown in figure 21a. A voltage divider creates a virtual ground at the center junction and draws less than 1 ma. at idle. The electrolytic ouput capacitors both reduce the supply impedance at high frequencies and function as a power reservoir to simulate two separate battery sources.

This version of a virtually grounded supply works best with amplifiers that draw lower idling current, as the capacitors must be able to “recharge” quickly. Start with 100uF capacitors. With an oscilloscope or a multimeter, monitor the supply for any ringing or fluctuation with the amplifier driving headphones loudly. Increase the capacitance or decrease the resistor values of the voltage divider to compensate (decreasing the voltage divider resistance values will increase idle current). The most important test of all is the listening test. Despite supply fluctations, the amplifier may function without audible detriment. For a simple AC power supply, apply this circuit to an adapter with 12V regulated output (Radio Shack sells one) to get regulated ±6V (figure 21c). Regardless of the supply option used, decoupling the opamps from the supply (figure 21d) will improve stability.

opamp20
Figure 22

There may be times, though, when a virtually-grounded dual supply has a tendency to “rail” when the resistor-type voltage divider cannot maintain the virtual ground at 1/2 Vcc. Such cases may occur when the opamp draws too much current or input signal (for example, a high boost equalizer) pushes the opamp into heavy clipping and power supply is unable to recover. There are several inexpensive commercial voltage references that can output a stable 1/2 Vcc regardless of the load. Figure 22 shows a virtually-grounded dual supply implemented with the Texas Instruments TLE2426 voltage reference and one 9V battery (the TLE2426 is excellent also with two 9V batteries for a stable, dual 9V supply). Before selecting a voltage reference, check the specifications to see if it has adequate current capability for the load.

Addendum

11/11/98: Expanded discussion of using opamps to drive headphones directly without an output stage. Also updated figures 21c, 21d – lowered voltage divider resistor values to 5K, based on experimentation.

11/13/98: Added image and description of pocket headphone amp.

5/10/99: Added the following new sections: Equalization, Acoustic Simulation, Adjusting Opamp DC-Offsets. Also, minor revisions in other sections.

7/19/99: Added section on headphone distribution amplifiers.

7/20/99: Updated section on headphone distribution amplifiers.

7/24/99: Updated section on headphone distribution amplifiers.

1/3/00: Updated the following sections: Selecting Solid State Opamps, Configuring Opamps for Voltage Gain, The Output Stage.

1/24/00: Corrected reversed opamp input connections in figure 18a and 18b. Also corrected calculation of Rs for MOSFET driver in figure 9.

4/10/00: Revised figure 16c and corresponding description.

12/10/07: Revised figure 9a. Added figure 9b and corresponding description. Thanks to PRR for the circuit.

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Also thanks to Richard Steven Walz for his insight on virtually-grounded dual power supplies.

c. 2001 Chu Moy.